Conducted EMI in PWM Inverter for Household Electric Appliance
Yo-Chan Son and Seung-Ki Sul
School of Electrical Engineering & Computer Science #024
Seoul National University
Kwanak P.O.Box 34, Seoul, Korea (ZIP 151-742)
http://eepel.snu.ac.kr e-mail: sulsk@plaza.snu.ac.kr
Abstract- This paper presents the characteristics of the con- Electrotechnical Standardization), etc [1]. For an example,
ducted EMI in the PWM inverter for household electric appli-
EN55 014 of CENELEC limits RF disturbances by electri-
ances and proposes a new active common-mode EMI filter. The
cal motor-operated appliances for household purposes [2].
conducted EMI spectrums are measured with changing the ac-
tive/passive filter stages and the common-mode EMI is extracted Power line filters are usually attached in front of the
from the total conducted EMI spectrums in order to identify the
equipment in order to suppress these side effects of the PWM
dominant source of the conducted EMI. For better understand-
inverter, but they result in additional cost and volume of the
ing of the conducted EMI, the noise sources are identified and
system. These filters are intended for the attenuation of the
the characteristics of the noise source and its propagation paths
unwanted high frequency signals, but their design is quite
are investigated from the conducted EMI spectrums measured.
difficult due to the identification of noise source and nonline-
Based on the measurement and analysis of the conducted EMI
arity of filtering elements at the high frequency region.
in the PWM inverter, the effective method and procedure of
Without better understanding of the noise source and the
designing filter stages are developed in order to attenuate the
conducted EMI effectively. characteristics of the system at the high frequency, the EMI
mitigation efforts could require excessive costs with degraded
I. INTRODUCTION
performance. In this paper, the conducted EMI spectrums of
the PWM inverter for household appliances are investigated.
As the progress of the power electronics technology, its
The implementation and modification of filter stages are de-
applications are being adopted not only in the industry, but
veloped and the common-mode EMI spectrum is extracted
also in the home appliances. The existing constant speed op-
from the total conducted EMI in order to separate the domi-
eration with on/off control is being replaced by the variable
nant noise source. Based on the measurement and analysis of
speed operation for better efficiency and quietness. For the
the conducted EMI in the PWM inverter, the effective meth-
operation of the ac motor PWM inverters are used to change
od of designing filter stages are developed in order to attenu-
the input frequency of the motor. Despite of these conven-
ate the conducted EMI effectively. A new active common-
iences, the radio frequency (RF) disturbance signal generated
mode EMI filter is introduced in order to increase the attenu-
by the PWM inverter makes its application be difficult. Large
ation performance of overall filter stages.
amount of high frequency noise is generated by the switching
operations of the PWM inverter. They are in form of con-
II. GENERATION AND PROPAGATION OF CONDUCTED EMI
ducted or radiated emissions. In case of the conducted emis-
sion, high frequency signals generated by the one equipments A. Fast dv/dt of Output Voltages
might interfere with other equipments that is connected at the
Fig. 1 shows the general structure of the PWM inverters for
point of common coupling (PCC), and invoke the malfunc-
single-phase ac input system. In order to meet the harmonic
tion of the victim. Also the radiated emission might disturb
regulations, the power factor correctors (PFC) are usually
the operation of wireless installations or add high frequency
used with a front-end single-phase diode-bridge rectifier [3].
conducted noise to the victim coupled with the conduction
Ac motors and compressors are used as load machine of
loops in the system. There has been a strong demand on the
PWM inverter. PWM inverters for motor drive applications
regulation of the electromagnetic interference (EMI) by
are operated at 1 ~ 20[kHz] of switching frequency and usu-
equipments in the customer electronics since 1990, and each
ally IGBTs are used as their power semiconductors switches.
of products should be compatible with national or interna-
Fast switching operations of IGBTs generate unwanted high
tional standards such as IEC (International Electrotechnical
frequency voltages and currents coupled with system para-
Commission), CENELEC (the European Committee for
3-Ć
Load
PCC
Ll
Cll
a
+
s
n
vAC
b
-
c
Csg
Clg
g
Fig. 1. PWM inverter connected to a single-phase ac input system.
Dashed line: path of normal-mode current, dotted line: path of common-mode current.
0-7803-7114-3/01/$10.00 (C) 2001
to-ground capacitances Clg and stator-to-ground capacitances
ig1 Csg. The high frequency current as shown in Fig. 2(a) is gen-
erated on the path indicated by dotted line in Fig. 1. As well
vsn
as the high frequency normal-mode current, the common-
(a) mode current is drawn from the ac power source and can be
emitted by loops made by safety earth wires or ground return
ig
paths. Especially loops for the common-mode current are
rather larger than those of the normal-mode current, which
act as good antennas for radiated EMI [4].
B. Inductive Load Current Switching
Fig. 2(b) shows the waveform of inverter input current
iinv when the induction motor or any other inductive load machi-
(b)
ias ne is connected in Fig. 1. The inverter input current iinv is de-
termined by the switching state of inverter as (3).
vsn
iinv = Saias + Sbibs + Scics
, (3)
where ias, ibs and ics are phase currents of the inverter. The
Fig. 2. High-frequency current generated by PWM inverter common-mode voltage vsn is depicted in Fig. 2 to represent
operations. (a) Leakage current by common-mode voltage the switching state of the inverter. During the active voltage
(Top: motor leakage current ig1, 200[mA/div]. Middle: applied in a PWM period, the inverter current follows the
common-mode voltage vsn, 100[V/div]. Bottom: total leak- phase current with respect to the switching state, and is
age current ig, 50[mA/div]. Time: 5[µs/div].), (b) Inductive clamped to zero when the zero voltage vector ([Sa, Sb, Sc] =
[0,0,0] or [1,1,1]) is applied in the PWM period. This inverter
load current switching (Solid line: inverter input current
input current waveform is analogous to that of the dc-dc Buck
iinv, 2.5[A/div]. Dashed line: a-phase current ias, 2.5[A/div].
converter, which has large harmonic contents of the high fre-
Common-mode voltage vsn, 100[V/div]. Time:
quency current. Large dc-bus capacitors are supposed to han-
200[µs/div].)
dle this high frequency current, and electrolytic capacitors are
used for that purpose. But they are relatively large effective
sitic components.
series resistance (ESR) and inductance (ESL), and have the
High frequency normal-mode currents are induced by the
limited high-frequency capability. Thus some portion of the
abrupt voltage transition of output line-to-line voltage cou-
high frequency inverter input current should be drawn from
pled with line-to-line capacitance Cll and stray inductances
the ac power source, and it can be a source of conducted EMI
such as Ll and some of them can be drawn from ac power
[5]. Unlike the dv/dt problems, this is strongly dependent
source as indicated by the dashed line in Fig. 1 if the imped-
upon the operating condition and the voltage modulation
ance of dc-bus capacitor is finite. This normal-mode current
method. It is well known that the rms value of the inverter
can be a source of conducted EMI, and also can be a source
input current remains same regardless of the modulation
of radiated EMI coupled with loops in output cables or power
method if the switching frequency of PWM inverter is much
stage layouts. Also switching operation of PFC can be a sour-
higher than that of output voltage and thus the magnitude of
ce of high-frequency normal-mode currents. Because the
the output current ripples can be negligible with respect to
semiconductor switch of the PFC is operated synchronously
that of the fundamental output current [6]. But the distribu-
with the line frequency only to maintain the continuous con-
tion of the frequency components of the inverter input current
duction of diode rectifier for harmonic reduction, the low-
in the frequency domain are clearly expected to be varied
frequency operation of PFC is expected not to be a major
with different modulation methods as in the case of [7].
source of high-frequency normal-mode current compared
III. MITIGATION OF CONDUCTED EMI
with others.
Fig. 3 shows the PWM inverter system considered in this
When it comes to common-mode EMI, the PWM inverter
paper. It is a compressor drive unit of an air conditioner, and
is the major source of the common-mode voltage, and its
its filter stages are modified in this paper for EMI analysis.
common-mode voltage can be defined by its switching con-
PFC in this system is operated at the ac line frequency. Its
dition. Except the operation of the PFC, the common-mode
switch is turned on at the moment of zero crossing point of
voltage in Fig. 1 can be represented as (1) and (2).
the input voltage, and remained as the on-state during some
Vdc Vdc
fixed time only for reduction of the harmonic pollution. Thus
vsn = (Sa + Sb + Sc )- (1)
the inductor in PFC should be very large in order to allow the
3 2
continuous conduction of input current, and 20[mH] of dc
and v = vsn + vng , (2)
sg
inductor with laminated silicon-steel E-I core is used in this
where Sa, Sb and Sc represent switching states of inverter
system. From the ac input terminal to dc-bus capacitor, all
bridges. vng changes slowly compared to the variation of vsn if
components including control circuits and SMPS circuits are
the operating frequency of PFC is quietly low, and most of
installed within a single PCB board and the IPM board for the
dv/dt comes from PWM switching operation in (1) as shown
IGBT inverter is connected to dc-bus capacitor in the PCB
in Fig. 2(a), where the system parameters and operating con-
board by wires for the convenience of the system layout. A
ditions are given in Table I. Fast dv/dt of the common-mode
snubber capacitor Cs is installed on the IPM board in order to
voltage excites parasitic components such as junction-to-
suppress the voltage spike at the power devices. A PFC
heatsink capacitances in power semiconductor switches, line-
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LISN 50&!-50µH
Inverter
Cx4 Cx3 Cx1
Motor
0.1µF
1µF
s
50&!
ACEF
50µH
LCM 2 Cy Cx 2 Cs
Fig. 3. Configuration of experimental system.
inductor and a 3.7[kW] induction motor are connected to PCB effects from components and system layout, which makes the
board by wires, and a ferrite bead (not included in Fig. 3) is high frequency modeling of the system be almost impossible.
installed on inverter output cables to the induction motor for There are several research results in which various active
the reduction of common-mode radiated EMI. filtering techniques are tried [9 ~ 12]. The purpose of the ac-
tive EMI filter is to provide greater attenuation of high fre-
EMI filters are usually located between the ac input termi-
quency signals under these limitations. Especially in case of
nal and the diode rectifier, and provide the attenuation of high
the active common-mode EMI filter, the signal to be sup-
frequency signals in order to suppress the conducted EMI
pressed is the relatively high frequency signal compared to
emission. In most cases, noise suppression is implemented
the normal-mode current ripple in SMPS applications [9, 10],
using LC low-pass filters, and the adjustment of the insertion
and the compensation circuit should be fast enough to elimi-
loss is done by changing their cutoff frequencies and in-
nate it without any phase delay.
creasing the number of filtering stages. But there are some
limitations in implementing filter stages. In case of normal- In this section, the variation of conducted EMI spectrum is
mode LC filter, the filter inductor should be able to handle investigated with respect to the variation of the filter stages.
the rated current of the system without the saturation of its In order to suppress the common-mode noise effectively, a
magnetic core. With this fact the core size should be in- new active common-mode EMI filter is introduced. Detailed
creased if the larger inductance is required, which are directly operating conditions and system parameters of the given sys-
related to the system cost and volume. The value of the nor- tem are listed in Table I. A LISN (Line Impedance Stabiliz-
mal-mode filter capacitor (X-capacitor) is also important in ing Network) is connected between the ac power source and
the low voltage system such as switched-mode power sup- the inverter system in order to provide a stable 50[&!] imped-
plies (SMPS), because the power factor of the system can be ance to the inverter system in frequency range of 150[kHz] ~
degraded by X-capacitor. But they are usually negligible in
30[MHz] [2, 8]. Measuring the total conducted EMI level of
high voltage system, and the adjustment of the normal-mode
the system, one of 50[&!] resistors in Fig. 3 is used as a dum-
insertion loss should be done by the increase of X-capacitor
my resistive load, and the other as the input impedance of the
without changing the filter inductor to some extent. There are
spectrum analyzer. The total conducted EMI is the sum of the
similar limitations in implementing the common-mode filter.
normal-mode EMI and common-mode EMI, and one cannot
The value of the common-mode filter capacitor (Y-capacitor)
tell the origin among different noise sources apart only with
is limited to several nF because of the safety purpose [8]. It is
the result of the total conducted EMI spectrum. A differential
possible to make the common-mode filter inductor provide
mode rejection network (DMRN) is used in order to measure
large inductance because the flux induced by the common-
the conducted common-mode noise separately as shown in
mode current is usually small. Thus the adjustment of the
Fig. 4 [13].
insertion loss is usually done by the increase of the common-
A. Active Common-mode EMI Filter (ACEF)
mode inductor size. Usually a common-mode inductor con-
tains not only the common-mode inductance but also the
Fig. 5 shows the proposed active common-mode EMI filter.
normal-mode inductance resulting from its leakage induc-
The proposed circuit is based on the ripple current elimina-
tance. Normal-mode filtering is also done by the common-
tion technique measuring the source current ripple [9 ~ 11].
mode inductance only with the aid of X-capacitor in small
power systems, but the separate normal-mode filter inductor
is required for greater attenuation in larger power systems.
LCM
Unlike the case of the filter design of signal processing appli-
cations, there are not only LC limitations mentioned above,
but also the difficulties in predicting the performance evalua-
tion of the designed filters because of unpredictable parasitic
BNC
16.7&!
rb
C0
LISN
50&! BNC
12V
Spectrum
16.7&!
Analyzer
16.7&!
LISN
50&!
Cc
300V
Fig. 4. Differential Mode Rejection Network (DMRN) for
C0
single-phase ac power source[13].
Fig. 5. Proposed active common-mode EMI filter(ACEF).
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(a)
(a)
(b)
(b)
(c)
(c)
Fig. 6. Conducted EMI when no EMI filter installed. (a)
Fig. 7. Conducted EMI when LCM is installed only. (a) To-
Total EMI, (b) Common-mode EMI, (c) Leakage cur-
tal EMI, (b) Common-mode EMI, (c) Leakage cur-
rent(200[mA/div], 5[ms/div]).
rent(200[mA/div], 5[µs/div])
In order to minimize the phase delay caused by the filter cir-
provide the low impedance path between the earth ground
cuit the single stage of the push-pull amplifier as in [11] and
and the output of the filter circuit while blocking the path of
[12] has been designed. The filter circuits in [11] and [12]
the low-frequency signal. Overall the series impedance of C0
utilizes the dc bus voltage as the power source of the push-
and Cc should be limited in order to keep the supply voltage
pull amplifier. In this case, both transistors should handle the
and to withstand the high voltage test.
full dc voltage, but the high voltage pnp transistor is hard to
obtain in the market and its manufacturing is not favorable as
B. Mitigation of Common-mode EMI
the dc bus voltage increases. The proposed circuit uses the
Fig. 6 shows the conducted EMI spectrum of the system
additional 12[V] voltage source that is used as the control
without any filtering elements between ac input terminal and
power supply in the inverter system. Due to the use of the
diode rectifier in Fig. 3. The peak detector is used in the EMI
low voltage power supply, it is possible to use faster devices
measurement [2, 8], and each waveform is video-averaged for
in the filter circuit and it can be applicable in any inverter
10 scans. The EMI characteristic above 10[MHz] strongly
application with any dc bus voltage compared to [11]. Also
relies on the experimental layout of the LISN and the EUT,
the proposed circuit does not care about the common-mode
and its filtering is not an easy job without the fundamental
voltage generated but tries to eliminate the common-mode
redesign of the system. Thus it will not be discussed in this
current compared to [12].
paper that focuses on the filtering technique. Both conducted
The source common-mode current makes the high frequen- EMI waveforms in Fig. 6(a) and (b) continuously droop be-
cy ripple flux in the common-mode choke depicted in Fig. 5,
tween the frequency bandwidth from 150[kHz] to 30[MHz].
which makes the high frequency voltage at an auxiliary
The waveform of the total EMI in Fig. 6(a) is slightly greater
winding. The high frequency voltage is then converted to the
than that of the common-mode EMI and one cannot tell that
high frequency current signal by the base resistor rb and the
which noise of the normal- and common-mode EMI is domi-
input impedance of the push-full amplifier. It excites the base
nant. The common-mode leakage current waveform is shown
current of the amplifier and then the amplifier provides the
in Fig. 6(c) that is directly returned to the source.
low impedance path for the high frequency signal between
A common-mode inductor LCM of Fig. 5 is installed in the
the inverter and the motor with respect to the sensed signal.
system of Fig. 6 without the push-pull amplifier circuit. As
In order to provide the high frequency current path between
can be seen in Fig. 7(a) and (b), the total EMI is also reduced,
the dc bus and the control power supply, two coupling ca-
but the reduction ratio is not same as that in the common-
pacitors C0 are used. These coupling capacitors should be
mode EMI because the leakage inductance of the common-
small enough to isolate the 12[V] power supply from the dc
mode inductor is just 7.5[µH]. With the increased impedance
link at low frequency. Thus the low-frequency common-
of the common-mode current path, the peak value of the
mode current is not suppressed with the proposed circuit,
leakage current is reduced as shown in Fig. 7(c) and also its
where its path is blocked by the coupling capacitor. Another
frequency is more sluggish than that in Fig. 6(c). The differ-
coupling capacitor Cc should be implemented in order to
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(a) (a)
(b)
(b)
Fig. 9. Conducted EMI when the additional passive filter
stage is added to the system of Fig. 8. (a) Total EMI, (b)
Common-mode EMI.
ig1
quency is around 800[kHz]. Because the ferrite core has the
(c)
high Q-factor, it makes the resonant peak around that fre-
ig
quency and the normal-mode EMI is boosted in that region.
With the aid of the ACEF and the additional passive com-
mon-mode filter stage the common-mode EMI is much re-
Fig. 8. Conducted EMI when ACEF is installed. (a) Total
duced but the total conducted EMI is not reduced to meet the
EMI, (b) Common-mode EMI, (c) Upper: motor leakage
limit line. The change of the total EMI spectrums shown in
current ig1, Lower: source leakage current ig. Fig. 6 ~ 9 shows that the dominant source of the conducted
EMI is in the form of the normal-mode EMI after the appro-
priate common-mode mitigation, and the strategy for the re-
ence between the total EMI and the common-mode EMI is
duction of the normal-mode noise is required in order to pro-
less than 10[dBµV] and is not enough to isolate the mitigation
ceed the filter design effectively.
effort of the normal-mode EMI.
C. Mitigation of Normal-mode EMI
Fig. 8 shows the effect of the proposed ACEF. High volt-
age capacitors of 10[nF] are used for output and coupling
When common-mode chokes are implemented in the pre-
capacitors. The push-pull amplifier circuit is added in the
vious section, two small normal-mode inductors are intro-
system of Fig. 7 and it suppresses the entire common-mode
duced as the leakage inductance of them. Thus it is possible
EMI waveform by 10[dBµV] at least. Fig. 8(b) shows that the
to build the multi-stage normal-mode filter. Total conducted
proposed circuit still provides fine attenuation until 10[MHz].
EMI spectrums according to the increase of the normal-mode
In Fig. 8(c), most of high-frequency signals in the returning
filter stages are shown in Fig. 10 ~ 12.
leakage current ig are disappeared by the proposed circuit,
At first Cx1 and Cx2 are added to the system of Fig. 9 in Fig.
and there is only some low frequency signal (less than
10. These capacitors are required in order to minimize the
100[kHz]) that cannot be suppressed by the circuit. Because
coupling of the common- and normal-mode noises caused by
the ACEF forces the motor leakage current ig1 kept in the
the circuit asymmetry of the PFC and also to keep the nor-
system and provides the low impedance path for that, the
mal-mode ripple current circulating within those capacitors.
circulating leakage current between the inverter and the mo-
Each capacitor is the polypropylene capacitor of 330[nF].
tor is slightly increased compared with that in Fig. 6 (c). The
The entire total EMI is much reduced due to the 2nd order
total EMI is also reduced and now the common-mode EMI is
low-pass filter formed by those X-capacitors and the leakage
clearly less than the total EMI and the dominant EMI source
inductance of common-mode chokes, and it is lower than the
is the normal-mode EMI. At this moment, the level of the
limit line almost frequency range except 1[MHz]. However
common-mode EMI is the ultimate low limit of the total EMI
there is little margin from the limit line below 1[MHz] and
if the appropriate filtering is provided for the normal-mode
the EMI level even exceeds at 1[MHz]. Thus more insertion
EMI. But the level of the common-mode EMI is still close to
loss is required in order to secure enough margins.
the limit line and more attenuation is required.
In order to construct the 4th order multistage filter for the
An additional passive filter stage is added to the system of
normal-mode EMI an additional X-capacitor Cx3 is inserted
Fig. 8. A common-mode inductor LCM2 and small Y-capacitors
between LCM and LCM2, and the total EMI waveform is meas-
Cy are used. After adding the additional common-mode filter,
ured as in Fig. 11. Unlike the expectation, the total EMI
the level of the common-mode EMI is quite less than the
waveform is not decreased at all compared with the result of
limit line in Fig. 9(b) but the total EMI at low frequency is
Fig. 10. This is because the source impedance, the LISN im-
slightly increased in Fig. 9(a) compared with that in Fig. 8(a).
pedance, is not so small compared to the input impedance of
It follows that the small leakage inductance of LCM2 with Y-
the normal-mode filter stage, and thus the filter stage cannot
capacitors forms a normal-mode filter which resonant fre-
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LISN Drive System Motor
+
s
EMI
Inverter
vAC
(a)
ig Filter
-
ig2 ig1
Fig. 10. Total conducted EMI with Cx1 and Cx2.
iCM 2 iCM - vcom +
s
+
2Cy
(b)
ACEF
vCM
- 0
3
1
2
iinv
+
Cx 4 Cs
Fig. 11. Total conducted EMI of multistage filter. (c) vDM
-
Cdc
Fig. 13. Propagation of conducted noise.
(a) Layout of measurement, (b) Propagation of common-
mode EMI, (c) Propagation of normal-mode EMI.
mode EMI voltage vCM across the 25[&!] resistor that is the
parallel resistance of the LISN. The proposed ACEF provides
the low impedance path 0 as indicated in Fig. 13(b) in order
Fig. 12. Final Result of total conducted EMI. to minimize iCM. The remaining high frequency current is
captured by the Y-capacitor. In this system the ACEF lowers
provide the enough attenuation. That is, the front-end imped-
the source leakage current, which keeps the additional com-
ance of this filter stage is 33[&!] at 150[kHz] (the leakage in-
mon-mode choke LCM2 from being saturated in case of the
ductance of LCM2), which is greater than the normal-mode
excessively large leakage current. Because the proposed cir-
impedance of the LISN, 100[&!]. In this case the additional X-
cuit is not working at the low frequency, the low frequency
capacitor should be provided near the source impedance in
leakage current is only dependent upon the Y-capacitor and
order to provide enough insertion loss with the mismatched
the system parasitic capacitance. That is, the proposed circuit
impedance condition. After adding the front-end X-capacitor
does not increase the low frequency leakage current that is
Cx4 the total EMI is much lower than the limit line and ap-
different from the behavior of the Y-capacitor. The additional
proaches closely to the ultimate limit of the attenuation, the
common-mode choke LCM2 blocks the high-frequency leakage
level of the common-mode EMI as shown in Fig. 12. Alt-
current that helps the internal circulation by the Y-capacitor
hough the additional Cx4 slightly increases the total EMI be-
(path 1 ) and ACEF (path 0 ). In the 2nd order low-pass fil-
tween 2 ~ 10[MHz], it does not matter because the total con-
ter formed by LCM2 and Cy, there can be some oscillatory cur-
ducted EMI in that region is pretty low enough.
rent of the low frequency, which results from the high Q fac-
tor of the ferrite core. It can be reduced by the use of the
Until now, the total EMI has been effectively suppressed
lossy material of the magnetic core but the inductance can be
with the reduction of the normal-mode EMI after reducing
lowered because the relative permeability is much less than
the common-mode EMI first. But experimental results reveal
that of ferrite. Or some proper damping method can be ap-
that additional filtering elements may not help the perfor-
plied such as in [14] and [15] without worsening the total
mance of existing filters as shown in Fig. 11. Moreover as the
attenuation capability.
resonant peak shown in Fig. 10 ~ 12, there can be some un-
expected results from the presence of parasitic components in
As far as the normal-mode EMI is concerned, the major
filtering elements and PCB layout, which makes the analysis
source of the normal-mode EMI is the input current of the
and application of filtering techniques difficult.
PWM inverter and its propagation of the normal-mode EMI
can be represented as shown in Fig. 13(c). The input current
IV. DISCUSSION
of the PWM inverter is highly pulsating and can contain more
Fig 13(a) shows the connection of earth wires in the given
high frequency components than the output current of the
system. A safety earth wire is connected at the earth terminal
PWM inverter. Some of high frequency normal-mode cur-
of the drive system including EMI filters, and the other wire
rents are absorbed by the snubber capacitor Cs that is attached
is routed to the earth terminal of the LISN. As mentioned
to the IPM board. But its attenuation performance is limited
before, the PWM inverter is the obvious common-mode volt-
by its capacitance, and this cannot be increased as the in-
age source of vcom, and the impulsive leakage current is gen-
crease of the capacitance of the dc-bus electrolytic capacitor.
erated as shown in Fig. 2(a), 6(c), 7(c) and 8(c). The sum of
The electrolytic capacitor has relatively large ESR and ESL,
currents in paths 2 and 3 in Fig 13(b) affects the common-
and is expected not to handle the high frequency normal-
0-7803-7114-3/01/$10.00 (C) 2001
mode EMI effectively. Thus most of high frequency normal- DC Link PWM Converter Systems, in IEE Conf. Rec. of Elec-
trical Machines and Drives, 1999, pp. 81 ~ 89.
mode currents should be blocked by external normal-mode
[7] L. Rossetto, S. Buso, and G. Spiazzi, "Conducted EMI Issues in
filtering elements. Stray inductances and capacitances can
a 600W Single-Phase Boost PFC Design," IEEE Trans. Ind. Ap-
make some unpredictable modes that worsen the attenuation
plicat., vol. 36, no. 2, Mar/Apr 2000, pp. 578 ~ 585.
performance of the normal-mode filter as shown in Fig. 10
[8] L. Tihanyi, Electromagnetic Compatibility in Power Electronics,
and Fig. 11. The 1[MHz] resonant peak has been made by the
IEEE Press, 1995.
resonance between the snubber capacitor Cs and the dc-bus
[9] L. Lawhite and M. F. Schlecht, Design of Active Ripple Filters
wire. The length of the total dc-bus wire is about 20[cm],
for Power Circuits Operating in the 1-10MHz Range, IEEE
which makes the stray inductance of 250[nH]. It is not pre- Trans. Power Electron., vol. 3, no. 3, Jul 1988, pp. 310-317.
[10] N. K. Poon, J. C. P. Liu, C. K. Tse and M. H. Pong, Tech-
sent in the common-mode EMI waveform as shown in Fig.
niques for Input Ripple Current Cancellation: Classification and
9(b) and can be suppressed only by the normal-mode filtering
Implementation, IEEE Trans. Power Electron., vol. 15, no. 6,
elements as shown in Fig. 12. If the coupling of the common-
Nov 2000, pp. 1144-1152.
mode choke is very high, which is the usual case, its leakage
[11] I. Takahashi, A. Ogata, H. Kanazawa, Active EMI Filter for
inductance can be rather smaller than that of the input imped-
Switching Noise of High Frequency Inverters, in Conf. Rec. of
ance of the system and the source impedance as in the case of
IEEE PCC-Nagaoka `97, 1997, pp. 331-334.
this system. Thus the desired attenuation cannot be obtained
[12] S. Ogasawara and H. Akagi, Circuit Configurations and Per-
because of the load effect of the filter. In order to minimize
formance of the Active Common-Noise Canceler for Reduction
this effect, appropriate X-capacitors should be installed in of Common-Mode Voltage Generated by Voltage-Source PWM
front of the source and the system [8, 15]. In this system, X- Inverter, in Conf. Rec. of IEEE IAS, 2000, pp. 1482-1488.
[13] M. J. Nave, A Novel Differential Mode Rejection Network for
capacitors, Cx2 and Cx4 are functioning in this manner. After
Conducted Emissions Diagnostics, IEEE National Symposium
minimizing the load effect the desired insertion loss can be
on Electromagnetic Compatibility, 1989.
obtained as in the case of Fig. 12.
[14] S. Ogasawara and H. Akagi, "Modeling and Damping of High-
Frequency Leakage Currents in PWM Inverter-Fed AC Motor
Another aspects to be considered is that the asymmetry of
Drive Systems," IEEE Trans. Ind. Applicat., vol. 32, no. 5,
the PFC may increase the interference between the common-
Sep./Oct. 1996, pp. 1105-1114.
mode EMI and the normal-mode EMI, which makes the filter
[15] M. J. Nave, Power Line Filter Design for Switched-Mode Pow-
design more difficult [7].
er Supplies, Van Nostrand Reinhold, 1991.
V. CONCLUSION
TABLE I.
SYSTEM PARAMETERS OF GIVEN SYSTEM.
In this paper, the characteristics of the conducted EMI of
Power rating Single-phase 220[V], 1.5[kW].
the PWM inverter for household appliances have been pre-
60[Hz] fixed output frequency,
sented and its mitigation efforts are introduced. A new active
PWM inverter discontinuous PWM with 2.5[kHz] switching
common-mode EMI filter is proposed in order to produce the
frequency.
effective insertion loss in the common-mode circuit without Switching device Mitsubishi DIP-IPM PS21205
the limitation of the passive filter. The conducted EMI spec- Load machine 3-phase 3.7[kW] induction motor
Line frequency switching
trums have been measured with changing and verifying the
PFC
for harmonic reduction.
effect of each filtering elements. The proposed circuit
LCM : 2[mH], N = 13, ferrite torroid,
provides good attenuation results without increasing the low-
7.5[µH] of leakage inductance
Common-mode choke
frequency leakage current. The spectrums of the common- LCM2 : 6[mH], N = 27, ferrite torroid,
35[µH] of leakage inductance
mode EMI have been separately measured in order to distin-
High voltage capacitor (2[kV])
guish the type of the dominant noise source, which enables
Y-capacitor and
Cy : 2.2[nF],
the contribution of the normal-mode EMI over the total EMI coupling capacitor
Cc and C0 : 10[nF]
to be examined. From these results the origin and character-
Polypropylene capacitor
istics of the noise source have been analyzed, and the cou- Cx1, Cx2 and Cx4 : 470[nF],
X-capacitor
Cx3 : 680[nF]
pling paths of the conducted EMI have been identified which
Cs : 100[nF]
leads to the way of the effective mitigation techniques of EMI.
Push-pull amplifier NEC 2SC3840 (pnp), 2SA1486 (npn)
REFERENCES
[1] T.Williams, EMC for Product Designers, 2nd ed., Newnes, 1996.
[2] EN 55 014 : 1993, Limits and Methods of Measurement of
Radio Disturbance Characteristics of Electrical Motor-operated
and Thermal Appliances for Household and Similar Purposes,
Electric Tools and Electric Apparatus
[3] L. Rossetto, P. Tenti and A. Zuccato, Electromagnetic Com-
patibility Issues in Industrial Equipment, IEEE Industry Appli-
cations Magazine, Nov./Dec. 1999, pp. 34 ~ 46.
[4] G. Skibinski, R. Kerkman, and D. Schlegel, "EMI Emissions of
Modern PWM ac Drives," IEEE Industry Applications Magazine,
Nov/Dec 1999, pp. 47 ~ 81.
[5] S. Chen, Generation and Suppression of Conducted EMI from
Inverter-Fed Motor Drives, in Conf. Rec. of IEEE IAS, 1999, pp.
1583 ~ 1589.
[6] J. Kolar, T. Wolbank and M. Schroedl, Analytical Calculation
of the RMS Current Stress on the DC Link Capacitor of Voltage
0-7803-7114-3/01/$10.00 (C) 2001
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