IEEE/ASME TRANSACTIONS ON MECHATRONICS, VOL. 6, NO. 2, JUNE 2001 111
Mechatronics of Electrostatic Microactuators for
Computer Disk Drive Dual-Stage Servo Systems
Yunfeng Li, Student Member, IEEE, and Roberto Horowitz, Member, IEEE
Abstract A decoupled control design structure and discrete slider, and it either rotates or translates the slider relative to
time pole placement design method are proposed for MEMS-based
the suspension. The actuated slider approach achieves a truly
dual-stage servo control design in magnetic disk drives. Dual-stage
collocated second stage actuation of the read/write head, by-
track following controllers are designed using a decoupled
passing nonlinear friction, bias forces, and all E-block, suspen-
three-step design process: the voice coil motor (VCM) loop design,
sion, and gimbal structural resonance modes. Usually, a MEMS
the microactuator (MA) inner loop design, and the MA outer loop
design. Both MIMO (when the MA relative position sensing is MA has a single flexure resonance mode in the 1 2-kHz fre-
available) and SIMO (when the MA relative position sensing is
quency range and has no other appreciable structural resonance
not available) designs are considered. The effect of MA resonance
modes up the 40-kHz frequency range [1], [3]. Thus, the MEMS
mode variations on the stability and performance of the controllers
MA actuated slider dual-stage approach may provide a potential
are analyzed. Self-tuning control and online identification of the
high-performance and low-cost solution to achieving extremely
MA model are developed to compensate for the variations in the
MA s resonance mode. high track density, since the MEMS MA can be batch-fabricated
and micro-assembled with the head and the gimbal of suspen-
Index Terms Dual stage, hard disk drive, servo control.
sion.
The resonance frequency of the MEMS MA s lightly damped
I. INTRODUCTION
flexure resonance mode is relatively low and close to the open
loop gain crossover frequency of the servo system. Furthermore,
T IS predicted that future areal storage density increases in
magnetic disk drive will be achieved mainly through an in- due to lithographic misalignment and variations present in the
I
crease in track density. For a predicted bit aspect ratio of 4:1, etching processes, the actual resonance frequency of the MA
an areal density of 100 Gb/in translates to a linear bit density can vary by as much as from its designed nominal value.
of 672k bits per inch (BPI), and a radial track density of 168k
Thus, the controller robustness to the uncertainty in the MA s
tracks per inch (TPI), which in turn implies a track pitch of 150
resonance frequency must be considered for MEMS-based dual-
nm. In order to achieve an ultimate tenfold increase in TPI, it
stage servo control design. Another difference between PZT
will be necessary to develop high-bandwidth, robust track-fol-
actuated suspension and the MEMS-based dual-stage system
lowing servo systems. Dual-stage actuation has been proposed
is the availability of the sensor that can measure the displace-
as a means of attaining the necessary servo bandwidth to achieve
ment of the read/write head, relative to the suspension. In most
the required runout and disturbances rejections.
PZT actuated suspensions, relative position sensing is gener-
Two dual-stage actuation approaches for magnetic disk drives
ally not available and the dual-stage controller must be single-
are currently being considered by the magnetic recording in-
input multi-output (SIMO), while, for most MEMS MAs, ca-
dustry. In the first approach, which is generally referred to as
pacitive or piezoresistive sensing can be used to measure the
the actuated suspension, mini-actuators (usually made of piezo-
MA s relative position [2], and the dual-stage controller can po-
electric materials such as PZT) are used to flex the suspen-
tentially be multi-input multi-output (MIMO).
sion around a pivot, producing relative motion of the read/write
Several controller design methods have been proposed for
head along the radial direction. However, most actuated suspen-
PZT actuated suspension dual-stage systems [4] [7]. Controller
sions have multiple structural resonance modes in the 4 12-kHz
designs for MEMS-based dual-stage servo systems have been
frequency range, which may limit the bandwidth of the servo
reported in [8] and [9]. In [8], a SIMO controller was designed
system. The second approach to dual-stage actuation utilizes
using a parallel design technique. However, the robustness of the
micro-electromechanical systems (MEMS) and is generally re-
servo system to variations in the microactuator resonance mode
ferred to as the actuated slider approach [1], [2]. In this ap-
was not addressed. In [9], robust SIMO and MIMO optimal
proach, an electrostatic or electromagnetic MEMS fabricated
controllers were designed using -synthesis. In this paper, we
microactuator (MA) is sandwiched between the gimbal and the
present a decoupled discrete time pole placement design method
which can be used for both SIMO and MIMO controller designs.
Manuscript received August 18, 2000; revised November 17, 2000. Recom-
The decoupling design approach utilized in this paper was orig-
mended by Guest Editors N. Matsui and M. Tomizuka. This work was supported
inally introduced in [10], for use in a PZT actuated suspension.
by the National Storage Industry Consortium (NSIC) and the Computer Me-
chanics Laboratory (CML) of University of California at Berkeley. Compared with the -synthesis design in [9], the implementa-
The authors are with the Department of Mechanical Engineering of the
tion of the controller designed using this method requires signif-
University of California at Berkeley, Berkeley, CA 94720-1740 USA (e-mail:
icantly less computations. Moreover, decoupled pole placement
horowitz@me.berkeley.edu).
Publisher Item Identifier S 1083-4435(01)03578-5. design can be combined with a self-tuning or an online
1083 4435/01$10.00 ©2001 IEEE
112 IEEE/ASME TRANSACTIONS ON MECHATRONICS, VOL. 6, NO. 2, JUNE 2001
TABLE I
IBM S ELECTROSTATIC MA PARAMETERS
the combined MA pico-slider assembly was designed to have
a nominal in-plane rotational flexural resonance frequency of
1.5 kHz. Variations in the etching process and lithographic mis-
alignments cause variations in the MA s resonance fre-
quency from its designed nominal value. Besides this single
lightly damped resonance mode, the MA does not have other ap-
Fig. 1. IBM s electrostatic MA mounted on an integrated lead suspension
preciable structural dynamics up to the 40-kHz frequency range.
(Courtesy of L.-S. Fan.).
Thus, its dynamics can be adequately described by a simple
mass-spring-damper second-order transfer function [1], [3]
identification scheme for compensating variations in the MA s
resonance frequency.
(1)
The paper is organized as follows. A MEMS-based dual-stage
actuator model is briefly introduced in Section II. In Section III,
the decoupled controller design structure and discrete time sen- Table I provides values for the parameters in (1) as well as
sitivity transfer function design process are presented. The ro- other important parameters of the IBM MA model, on which
bustness of the designs is analyzed in Section IV using -anal- the designs in this paper are based.
ysis. In Section V, MA inner-loop self-tuning control and online A rotational MA design was selected, in order to counteract
identification of the MA model parameters for compensating the the hundreds of in plane acceleration that is exerted to the
MA s resonance mode variations are discussed. Conclusions are pico-slider s center of mass by the voice coil motor (VCM)
provided in Section VI. during a seek operation. The pico-slider is mounted on the MA
in a way that its center of mass coincides with the actuator s
axis of in-plane rotation. Considering VCM as a rigid body,1
II. DUAL-STAGE MODEL WITH A MEMS MA
the equations of motion of the dual-stage system are
Fan and co-workers at the IBM Almaden Research Center
(2)
(ARC) have developed an innovative etching and electro-
plating multilayer technology for fabricating high-aspect-ratio (3)
MEMS devices [1]. This fabrication process can be imple-
where
mented at IBM s existing head manufacturing facilities, which
annually produce hundreds of millions of thin-film heads. (4)
The process includes high-aspect-ratio transformer-coupled
and and are, respectively, the moment of inertia of the
plasma (TCP) reactive-ion etching (RIE), which achieves a
VCM and the MA, is the mass of the MA, is the distance
20:1 height-to-width aspect ratio on 40-mm-thick polymers.
between the mass center of the MA and the pivot of the VCM,
The etched polymers are in turn used as molds to fabricate,
is the angular position of the VCM, is the angular po-
through a metal electroplating process, high-aspect-ratio invar
sition of the MA relative to the VCM, is the torque input
(a nickel iron-based alloy) micro-structures. Area-efficient
to the VCM, is the torque produced by the MA, is the
electrostatic rotational MAs have been successfully designed
damping coefficient of the MA, and is the stiffness of the
and fabricated using this process. Fig. 1 shows a photograph of
MA. Given the fact that the inertia of the MEMS MA is very
such a device mounted on an integrated lead suspension.
small compared to that of the VCM, for
A pico-slider is attached on top of the MA. Electrical contacts
the model used in this paper, we can neglect the coupling term
between the slider and MA are made using laser reflow. The
in (2) and assume that motion of the MA has no effect
electrostatic MA is assembled on to the gimbal of an integrated
on the motion of the VCM. Equation(3) can be rewritten as
lead suspension and rotates the pico-slider about its center of
(5)
mass, using an area-efficient layout of electrostatically actuated
comb-fingers.
where
Since the pico-slider flies on top of the disk on an air bearing,
(6)
the MA must support a gram of out-of-plane loading, with min-
imum out-of-plane deflection. Thus, the MA s flexures must be
Equation (6) shows that the motion of the MA can be decou-
very stiff in the out-of-plane direction while compliant in the
pled from the VCM by feeding to the MA with a proper
in-plane radial direction in order to have enough dc gain. A
1
Through this paper, we will refer to the combined VCM, E-block, and sus-
400:1 out-of-plane/in-plane stiffness ratio was achieved by the
pension assembly as VCM. The assumption that the VCM is a rigid body in
20:1 height width flexure aspect ratio in the fabrication process
this section is to simplify the dynamics analysis. In the subsequent section, it
described above. Because of out-of-plane deflection constraints, includes the resonance modes of the E-block and suspension.
LI AND HOROWITZ: MECHATRONICS OF ELECTROSTATIC MICROACTUATORS FOR DUAL-STAGE SERVO SYSTEMS 113
Fig. 2. Dual-stage block diagram.
Fig. 3. Dual-stage control design block diagram.
gain. As a consequence, for controller design purposes, the dual-
stage system can be approximated by the block diagram shown
in Fig. 2, where and are, respectively, the VCM and
MA transfer functions (TF), is the absolute position of the
read/write head, and is the absolute position of the tip of the
suspension). Thus, is the summation of and the position
of the MA relative to the suspension, which will be referred to
in this paper as , and is defined as
Fig. 4. The sensitivity block diagram.
(7)
The controller design in this paper is based on this dual-stage
In the block diagram shown in Fig. 3, there are three com-
model. For implementation, the actual control torque to the MA
pensators that need to be designed: the VCM loop compensator
is given by
, the MA loop compensator , and the MA
minor loop compensator . is
(8)
used to damp the MA s flexure resonance mode and place the
closed-loop poles of the MA loop at an appropriate
where and are generated from the designed controller.
location. The damped MA closed-loop transfer function ,
Capacitive sensing can be used in MEMS electrostatic MAs
shown in the lower-middle dashed box, is defined as
to measure the [2]. However, this requires additional
(10)
sensing electronics and wires to and from the head gimbal as-
sembly (HGA), which may result in an unacceptable increase
while the total dual-stage open loop TF from to , , is
in the fabrication and assembly costs. Thus, whether or not the
given by
will be used in MEMS dual-stage servo systems is still
(11)
an open question. In this paper, we will classify dual-stage track-
The block diagram in Fig. 3 is equivalent to the sensitivity
following controllers into two categories, according to the avail-
block diagram shown in Fig. 4, and the total closed-loop sensi-
ability of the : those utilizing the will be called
tivity TF from to is the product of the VCM and MA
multi-input multi-output (MIMO) controllers, while those not
loop sensitivity TFs, respectively, and :
utilizing the will be called single-input multi-output
(SIMO) controllers.
(12)
where
III. DECOUPLED TRACK-FOLLOWING CONTROLLER DESIGN
(13)
The block diagram for a MIMO decoupling control design
proposed in this paper is shown in Fig. 3. Thus, the dual-tage servo control design can be decoupled
The part enclosed in the dashed box on the upper-right corner into two independent designs: the VCM loop, whose error re-
of Fig. 3 is the open loop system described in Fig. 2. represents jection loop sensitivity TF is given by in (13), and the MA
the track runout, is position error signal of the head rela- loop, whose error rejection loop sensitivity TF is given by
tive to the data track center (i.e., ), while in (13).
is the position error signal of the head relative to the tip of the
A. VCM Closed-Loop Sensitivity
suspension (i.e., ).
The decoupling control approach, originally introduced by
The VCM loop compensator is designed to attain a de-
[10], utilizes the and to generate the position error
sired VCM closed loop sensitivity . Its bandwidth is gen-
of the suspension tip relative to the data track center, which will
erally limited by the E-block and suspension resonance modes.
be labeled as
The design of this compensator can be accomplished using con-
ventional SISO frequency shaping techniques and will not be
(9)
discussed here in detail. For illustration purposes, a sixth-order
and this signal is fed to the VCM loop compensator.2
model of the VCM, which includes torsional and sway vibration
modes of the suspension at 2.4 and 5 kHz, respectively, was uti-
2
In [10], the RPES was not assumed to be available and was estimated with
an open loop observer gain. lized in our simulation study and a fourth-order compensator
114 IEEE/ASME TRANSACTIONS ON MECHATRONICS, VOL. 6, NO. 2, JUNE 2001
Fig. 5. VCM loop sensitivity S magnitude bode plot.
Fig. 6. MA loop sensitivity S magnitude Bode plot.
running at a 20-kHz sampling rate was designed to have an
minor loop polynomials and are
approximated bandwidth of Hz. The gain Bode plot
both first-order. The resulting closed-loop transfer function
of the resulting sensitivity TF is shown in Fig. 5.
is given by
B. MA Closed-Loop Sensitivity
(22)
The MA loop controller is designed to increase the overall
closed-loop sensitivity attenuation. This design process is ex-
2) MA Outer Loop Controller Design by Pole Place-
plained in detailed in Section III-C, after we review the pole
ment: Consider now the design of the MA outer loop
placement design method applied to the MEMS MA.
compensator and as-
1) MA Minor Loop Controller Design by Pole Place-
sume that the closed-loop transfer function is given
ment: The zeroth-order-hold discrete time transfer function for
by (22). The resulting MA closed-loop sensitivity transfer
the MA model given in (1) is
function , which was defined in (13), is given by
(14)
(23)
where is a one-step delay operator and and
where the MA outer loop closed-loop polynomial
are, respectively, the MA open loop zero and pole
must be chosen by the designer. The MA outer loop compen-
polynomials
sator can be designed by
(15) solving the following Diophantine equation:
(16)
(24)
(17)
As in Section III-B1, it is convenient to define the second-
order polynomial in terms of the equivalent contin-
uous time natural frequency and damping ratio
(18)
and normally .
(19)
For , the low-frequency attenuation of
is given by
(20)
where is the controller sampling time and and
(25)
are, respectively, the MA s natural frequency and damping ratio,
given by Table I.
where is the open loop MA zero given by (18),
Consider now the minor loop feedback system en-
, , and is the sam-
closed by the dashed box in the lower part of Fig. 3 and the re-
pling time. Notice that . Thus,
sulting MA closed-loop TF defined by (10). Its poles
the ratio can be used to roughly determine the
can be placed by solving the following Diophantine equation
increased attenuation provided by the MA.
[11]:
The gain Bode plot of the sensitivity TF is shown in
Fig. 6, for the case when kHz, 300 MHz,
, and a 20-kHz sampling rate.
(21)
The closed-loop polynomial in (21) is chosen by the
C. Dual-Stage Closed-Loop Sensitivity
designer and its roots are the damped MA poles. It is convenient
to define the second-order polynomial in an analo- The MIMO dual-stage servo system depicted in Fig. 3 can
gous manner to the open loop polynomial in (16), in be designed by a three-step design process, which is based on
terms of the equivalent continuous time natural frequency (12) (25), and is schematically illustrated by Fig. 7.
and damping ratio , (i.e., substitute and , respec- First, the VCM compensator is designed to attain a
tively, by and in (17) (20)). desired VCM closed-loop sensitivity . In a typical design,
is a design parameter which will be specified later on the VCM bandwidth in Fig. 7 is limited by the existence
and normally . When is second order, the of E-block and suspension resonance modes to be between
LI AND HOROWITZ: MECHATRONICS OF ELECTROSTATIC MICROACTUATORS FOR DUAL-STAGE SERVO SYSTEMS 115
Fig. 8. SIMO control design block diagram.
results in an unnecessary decrease in attenuation . We pro-
posed that be initially chosen to be the same as . It
can subsequently be adjusted so that the desired attenuation and
phase margin requirements of the overall dual-stage system are
met. Decreasing increases the low-frequency attenuation
of . However, this generally reduces the phase margin of the
overall open loop transfer function in (11).
D. SIMO Track-Following Controller Design
The design procedure describe above can also be applied to
SIMO control architectures, where the is not available,
by incorporating an open loop observer to estimate the ,
as shown in Fig. 8. In this case, the open loop observer , as
defined in (22), generates the estimate signal .
By (21), the combined action of the open loop observer
and the minor loop compensator is equiva-
Fig. 7. Illustration of dual-stage sensitivity S design.
lent to the notch filter, , which is defined by
400 700 Hz. Subsequently, the MA compensators are designed
(26)
to attain the additional attenuation, , provided by the MA
closed-loop sensitivity . As discussed in Section III-B, this
where is the nominal MA open loop pole polynomial,
is accomplished by a two-step process. First, the minor
and is the desired pole polynomial of .
loop compensators, , are designed in order to damp
Ideally, if , the notch filter cancels the
the MA resonance mode and place the poles of , at a desired
MA open loop poles and assigns the poles defined by .
location. This involves the solution of the Diophantine (24).
However, the stability of the system is very sensitive to the un-
Since the poles of become zeros of , this is equivalent to
certainty in the lightly damped MA resonance frequency, .
placing the first corner frequency of in Fig. 7. Finally,
In order to make the notch filter more robust to the vari-
the loop compensator is designed to determine the
ations in , a wider notch filter can be used by setting a
overall bandwidth of the MA loop closed-loop sensitivity
larger damping ratio for . However, this will degrade
. This can also be achieved by pole placement or other
the performance of system, as shown in the step response in the
loop-shaping techniques.
next section. A better method for compensating the variations in
is limited by the sampling frequency, which is
the MA resonance frequency is to identify the MA poles online
assumed to be 20 kHz in this paper, and the controller computa-
and then generate the controller using the identified model. In
tional time delay, which was estimated to be 1/3 the controller
Section V-B, we introduce an online identification method that
sampling time. In [13], guidelines were developed for deter-
does not require the sensing.
mining the allowable servo bandwidth based on the available
sampling time and the computational delay time and suggest a
E. Design and Simulation Results
bandwidth of approximately 2.5 kHz for the above men-
In this section, we show simulation results of the proposed
tioned sample rate and time delay.
MIMO and SIMO designs, when the sampling frequency
The total dual-stage sensitivity is schematically shown in
is 20 kHz. Fig. 9 shows the gain Bode plot of the closed-loop
the bottom part of Fig. 7. For a given PES sampling rate and
time delay, and thus the bandwidth , the additional atten- sensitivity TF from track runout to , for both the MIMO
and SIMO designs. The design parameters used in the simula-
uation provided by the MA loop will be determined by the
selection of the lower corner frequency . Since se- tions were 500 Hz, 300 Hz, ,
lecting to be larger than the VCM loop bandwidth, 3000 Hz, and . In the case of the MIMO design,
116 IEEE/ASME TRANSACTIONS ON MECHATRONICS, VOL. 6, NO. 2, JUNE 2001
Fig. 9. S and S magnitude Bode plot.
Fig. 10. Step response of MIMO design.
is fourth order, both and are first order. The and SIMO designs. The SIMO controller response has a larger
gain crossover frequency (GCF), gain margin (GM), and phase overshoot and exhibits more residual vibrations than the MIMO
margin (PM) of open loop TF from to are, respectively, controller response. This is due to the fact that the MA s reso-
2337 Hz, 9.1 dB, and . In the case of the SIMO design, the nance poles are not being exactly canceled by the notch filter
combined VCM loop compensator is fifth order, and the com- in (26), in order to guarantee the stability robustness of
bined MA loop compensator is fourth order. When the damping the designed controller to MA resonance frequency variations.
ratio of was set to be 10 times larger than the MA TF, Notice that the settling time of the step response is rela-
the GCF, GM, and PM of open loop TF from to are, re- tively long. A decoupled dual-stage feedforward control can
spectively, 2432 Hz, 8.7 dB, and . Figs. 10 and 11 respec- be used to reduce the overshoot and settling time for track
seeking [12].
tively show the 1- m time-domain step responses of the MIMO
LI AND HOROWITZ: MECHATRONICS OF ELECTROSTATIC MICROACTUATORS FOR DUAL-STAGE SERVO SYSTEMS 117
Fig. 11. Step response of SIMO design.
IV. ROBUST STABILITY ANALYSIS USING
The robustness of the decoupled MIMO and SIMO designs
presented in Section III can be analyzed using the -analysis
[14]. , the structured singular value, is a measure of how
big a perturbation to a system must be in order to make the
closed-loop system unstable. For the case of MEMS-based
dual-stage servo systems, we are particularly concerned with
the robustness of the closed-loop system to variations in the
resonance frequency (or equivalently the stiffness) of the MA,
Fig. 12. -synthesis design block diagram.
unmodeled high-frequency structural resonance modes of the
VCM actuator, and variations in the dc gain of the VCM. In
the robust stability analysis framework, model uncertainties
a peak value of for the SIMO design. Thus,
are represented using linear fractional transformations (LTF).
both designs can maintain stability for the structured uncertain-
Fig. 12 shows the block diagram that was used to describe
ties described above and the MIMO design has better stability
structured uncertainties in our system. Three model uncer-
robustness than the SIMO design, as expected.
tainties , , and are considered in Fig. 12: represents
Of these three structured uncertainties, it is the variations in
the additive uncertainty used to describe the VCM unmodeled
the MA s stiffness which appear to have the most significant
resonance dynamics and is the frequency-shaped weight
detrimental effect on the performance of both the MIMO and
for . An uncertainty with the size of the amplitude of the
SIMO designs, even when the closed-loop system remains
biggest resonance peak of the VCM model was used for .
stable. For the MIMO design, variations in the stiffness of
is the parameter uncertainty that represents VCM loop gain
the MA s flexures produce variations in its dc gain. These dc
variations. is the parameter uncertainty that represents MA
gain variations have a large effect on the gain of the open loop
stiffness variations. We assume that both the VCM dc gain and
transfer function , which is dominated at low frequencies by
the MA resonance frequency can change by 10% from their
the last term of (11). For the SIMO design, the lightly damped
respective nominal values.
MA resonance mode can be excited if there is a significant
Fig. 13 shows the resulting robust stability plots. It shows mismatch between the actual MA resonance frequency and its
a peak value of for the MIMO design and nominal value.
118 IEEE/ASME TRANSACTIONS ON MECHATRONICS, VOL. 6, NO. 2, JUNE 2001
Fig. 13. Stability plot.
that will be presented is based on the pole placement design and
requires measurement of the MA s relative position error signal
.
Consider the MA open loop transfer function in (14). Since
the MA s resonance mode is lightly damped, the zero in (15)
and (18) remains fairly invariant with variations in the
resonance frequency and, moreover, . Thus, it is
possible to factor out the known term from the
Diophantine equation (21).
The resulting minor-loop closed-loop dynamics is
given by
Fig. 14. Self-tuning control of the MA.
V. COMPENSATION OF THE VARIATIONS IN THE MA S
(27)
RESONANCE MODE
where is the control input to the MA and denotes the po-
Two adaptive schemes are developed for compensating the
sition of the MA relative to the VCM, i.e., . Defining
variations in the MA s resonance mode. The first one utilizes a
direct self-tuning algorithm to tune the MA inner loop
(28)
controller, which can be applied to the MIMO design. For the
(29)
second method, the open loop MA model parameters are first
the regressor vector and filtered regressor vector as
identified, and the control parameters are subsequently com-
puted with pole placement based on the identified MA param-
eters. This scheme is applicable to both the MIMO and SIMO
(30)
designs.
(31)
A. Inner Loop Self-Tuning Control and the controller parameter vector , the
closed-loop dynamics (27) can be rewritten as
The block diagram for the MA inner loop self-tuning control
is shown in Fig. 14. The parameter adaptation algorithm (PAA) (32)
LI AND HOROWITZ: MECHATRONICS OF ELECTROSTATIC MICROACTUATORS FOR DUAL-STAGE SERVO SYSTEMS 119
Fig. 15. Control parameters adaptation response.
From (32), the controller parameter vector estimate
can be updated using a standard re-
cursive least square algorithm (RLS) [11]
(33)
(34)
Fig. 16. MA model identification with no RPES sensing.
Fig. 15 shows the simulation of the controller parameter esti-
mates , , , and , for the case when the real
(35)
MA resonance frequency is 1.2 times its nominal value, and the
system was subjected to stochastic disturbances, as discussed
The control law is
above. The controller parameters converged to a value, which
(36)
is very close to their desired value. Similar responses were
obtained when the real resonance frequency is 0.8 times the
with
nominal value. If the noise level is too large, the control
(37)
parameter estimates may not converge to their desired values
(38)
and other PAA s, such as normalized or extended RLS, can be
used for better convergence [11].
and the output of the MA fixed outer loop compensator
and the control input to the MA.
B. Online MA Model Identification
Fig. 15 shows the simulation of the control parameters
estimates using the RLS algorithm. In the simulation, realistic Since the variations in the MA s resonance frequency
estimates of the runout, VCM and MA torque disturbances, are due to its fabrication process, and does not change
and and measurement noises were injected into after the MA has been fabricated, it is feasible to identify the
the dual-stage system at corresponding locations. A white MA open loop dynamics online during the drive s manufac-
noise with an rms value of 10 nm was injected to the turing and testing stage. The controller parameters can then be
sensing signal. This noise is mainly due to thermal noise and subsequently determined, based on the identified MA plant pa-
feed-through in the capacitive sensing electronics [2], [15]. rameters, by solving (21) and (24).
120 IEEE/ASME TRANSACTIONS ON MECHATRONICS, VOL. 6, NO. 2, JUNE 2001
Fig. 17. Control parameters adaptation responses.
When the is available, the VCM feedback loop can effect of runout, VCM and MA torque disturbances, and
be closed using the as defined in (9). The open loop MA measurement noise.
dynamics can be identified by feeding an excitation input signal Fig. 17 shows the simulation of the MA parameters estimates
of a sufficiently large magnitude to the MA. Since the decou- , , , and using the extended recursive least square
pling control structure of the dual-stage servo system prevents (ERLS) algorithm [11]. In the simulation, realistic estimates of
the VCM motion from exciting the MA dynamics, the identifi- the runout, VCM and MA torque disturbances, and mea-
cation of the MA open loop model parameters can be carried out surement noise were injected into the dual-stage system at cor-
using standard identification techniques, based on the model responding locations. was a white excitation with a suffi-
ciently large amplitude to generate about m of MA motion
(39)
and was chosen to be fourth order. As shown in Fig. 17,
where is the input excitation signal to the MA, ,
the parameters estimates converged to their true values.
are its open loop pole and zero polynomials, as defined in (14),
is and is measurement noise.
VI. CONCLUSION
When sensor is not available and the only available
MIMO and SIMO track-following controllers for MEMS-
feedback signal is the , it is still possible to identify the MA
based dual-stage servo systems were designed using a decou-
open loop dynamics by closing the VCM loop with , and
feeding an excitation input signal of a sufficiently large magni- pled discrete time pole placement design methodology. Both de-
signs are robust to variations in the MA s resonance frequency.
tude to the MA, as shown in Fig. 16.
The MIMO design can achieve a superior robustness and per-
Defining this time to be the , we obtain
formance, partly due to the additional sensor. The de-
(40)
coupled MIMO design presented in this paper requires consid-
(41)
erable fewer computations than the synthesis design in [9].
where is the input excitation signal to the MA, is
Self-tuning control, or online estimation of the MA model pa-
the VCM loop sensitivity TF defined in (13), is the runout
rameters combined with the pole placement design, can be used
and is MA TF. By the spectral factorization theorem, the
to compensate for the variations in the microactuator s reso-
dynamics in (40) can modeled by
nance frequency, and restore nominal controller performance.
(42)
ACKNOWLEDGMENT
where , are the MA open loop pole and zero polynomials,
which need to be identified, is a fictitious white noise, The authors thank L.-S. Fan and W.-M. Lu from IBM, M.
and the Hurwitz polynomial represents the combined Kobayashi from Hitachi, and the industrial participants of the
LI AND HOROWITZ: MECHATRONICS OF ELECTROSTATIC MICROACTUATORS FOR DUAL-STAGE SERVO SYSTEMS 121
NSIC EHDR servo team for their comments and useful discus- [12] M. Kobayashi, T. Yamaguchi, and R. Horowitz, Track-seeking con-
troller design for dual-stage actuator in megnatic disk drives, in Proc.
sions.
Amer. Automatic Control Conf., June 2000, pp. 2610 2614.
[13] M. T. Whilte and W. -M Lu, Hard disk drive bandwidth limitations
due to sampling frequency and computational delay, in Proc. 1999
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Yunfeng Li (S 01) received the B.S. and M.S. de-
S. Chan, R. Horowitz, S. Aggarwal, and D. A. Horsley, High-Band-
grees from Beijing University of Aeronautics and As-
width High-Accuracy Rotary Microactuators for Magnetic Disk Drive
tronautics, Beijing, China, in 1992 and 1995, respec-
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gree in the Department of Mechanical Engineering,
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Roberto Horowitz (M 89) was born in Caracas,
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Venezuela, in 1955. He received the B.S. degree
dual-stage controller design and implementation for high track density
with highest honors in 1978 and the Ph.D. degree in
HDD s, in Proc. Amer. Control Conf., June 1999, pp. 4111 4115.
1983 in mechanical engineering from the University
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of California at Berkeley.
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In 1982, he joined the Department of Mechanical
2271 2273, Sept. 1999.
Engineering at the University of California at
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Berkeley, where he is currently a Professor. He
track-following servo design for hard disk drive, in Proc. Amer. Control
teaches and conducts research in the areas of adap-
Conf., June 1999, pp. 4188 4121.
tive, learning, nonlinear and optimal control, with
[10] K. Mori, T. Munemoto, H. Otsuki, Y. Yamaguchi, and K. Akagi, A
applications to micro-electromechanical systems
dual-stage magnetic disk drive actuator using a piezoelectric device for
(MEMS), computer disk file systems, robotics, mechatronics and intelligent
a high track density, IEEE Trans. Magnetics, vol. 27, pp. 5298 5300,
vehicle and highway systems (IVHS).
Nov. 1991.
Dr. Horowitz was a recipient of a 1984 IBM Young Faculty Development
[11] K. J. Åström and B. Wittenmark, Computer Controlled Systems: Theory
Award and a 1987 National Science Foundation Presidential Young Investigator
and Design. Englewood Cliffs, NJ: Prentice-Hall, 1984.
Award. He is a member of ASME.
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