SMPSRM/D
Rev. 3, Jul-2002
SWITCHMODE"! Power
Supply Reference Manual
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SRM/D
SWITCHMODEt Power Supplies
Reference Manual and Design Guide
SMPSRM/D
Rev. 3, July 2002
SCILLC, 2002
Previous Edition 2000
All Rights Reserved
SMPSRM
ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make
changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any
particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all
liability, including without limitation special, consequential or incidental damages. Typical parameters which may be provided in SCILLC data sheets and/or
specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including Typicals must be
validated for each customer application by customer s technical experts. SCILLC does not convey any license under its patent rights nor the rights of others.
SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications
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SMPSRM
Forward
Every new electronic product, except those that are battery powered, requires converting off line
115 Vac or 230 Vac power to some dc voltage for powering the electronics. The availability of design
and application information and highly integrated semiconductor control ICs for switching power
supplies allows the designer to complete this portion of the system design quickly and easily.
Whether you are an experienced power supply designer, designing your first switching power
supply or responsible for a make or buy decision for power supplies, the variety of information
in the SWITCHMODE Power Supplies Reference Manual and Design Guide should prove
useful.
ON Semiconductor has been a key supplier of semiconductor products for switching power supplies
since we introduced bipolar power transistors and rectifiers designed specifically for switching
power supplies in the mid 70 s. We identified these as SWITCHMODE products. A switching
power supply designed using ON Semiconductor components can rightfully be called a
SWITCHMODE power supply or SMPS.
This brochure contains useful background information on switching power supplies for those who
want to have more meaningful discussions and are not necessarily experts on power supplies. It also
provides real SMPS examples, and identifies several application notes and additional design
resources available from ON Semiconductor, as well as helpful books available from various
publishers and useful web sites for those who are experts and want to increase their expertise. An
extensive list and brief description of analog ICs, power transistors, rectifiers and other discrete
components available from ON Semiconductor for designing a SMPS are also provided. This
includes our newest GreenLine, Easy Switcher and very high voltage ICs (VHVICs), as well as
high efficiency HDTMOS and HVTMOS power FETs, and a wide choice of discrete products
in surface mount packages.
For the latest updates and additional information on analog and discrete products for power supply and
power management applications, please visit our website: (http://onsemi.com).
MEGAHERTZ, POWERTAP, SENSEFET, SWITCHMODE, and TMOS are trademarks of Semiconductor Components Industries,
LLC. HDTMOS and HVTMOS are registered trademarks of Semiconductor Components Industries, LLC.
GreenLine, SMARTMOS and Motorola are trademarks of Motorola Inc.
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3
SMPSRM
Table of Contents
Page
Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Linear versus Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Switching Power Supply Fundamentals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
The Forward Mode Converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
The Flyback Mode Converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Common Switching Power Supply Topologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
Interleaved Multiphase Converters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Selecting the Method of Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
The Choice of Semiconductors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Power Switches . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
The Bipolar Power Transistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
The Power MOSFET . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Driving MOSFETs in Switching Power Supply Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
The Insulated Gate Bipolar Transistor (IGBT) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Rectifiers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
The Magnetic Components . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Laying Out the Printed Circuit Board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Losses and Stresses in Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Techniques to Improve Efficiency in Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
The Synchronous Rectifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Snubbers and Clamps . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
The Lossless Snubber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
The Active Clamp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Quasi Resonant Topologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Power Factor Correction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
SMPS Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Integrated Circuits for Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Suggested Components for Specific Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
Literature Available from ON Semiconductor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56
Application Notes, Brochures, Device Data Books and Device Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56
References for Switching Power Supply Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58
Books . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58
Websites . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59
Analog ICs for SWITCHMODE Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61
ON Semiconductor Worldwide Sales Offices . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62
ON Semiconductor Standard Document Type Definitions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63
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SMPSRM
A low drop out (LDO) regulator uses an improved
Introduction
output stage that can reduce Vdrop to considerably less
The never ending drive towards smaller and lighter
than 1.0 V. This increases the efficiency and allows the
products poses severe challenges for the power supply
linear regulator to be used in higher power applications.
designer. In particular, disposing of excess heat
Designing with a linear regulator is simple and cheap,
generated by power semiconductors is becoming more
requiring few external components. A linear design is
and more difficult. Consequently it is important that the
considerably quieter than a switcher since there is no
power supply be as small and as efficient as possible, and
high frequency switching noise.
over the years power supply engineers have responded to
Switching power supplies operate by rapidly switching
these challenges by steadily reducing the size and
the pass units between two efficient operating states:
improving the efficiency of their designs.
cutoff, where there is a high voltage across the pass unit
Switching power supplies offer not only higher
but no current flow; and saturation, where there is a high
efficiencies but also greater flexibility to the designer.
current through the pass unit but at a very small voltage
Recent advances in semiconductor, magnetic and passive
drop. Essentially, the semiconductor power switch
technologies make the switching power supply an ever
creates an AC voltage from the input DC voltage. This
more popular choice in the power conversion arena.
AC voltage can then be stepped up or down by
This guide is designed to give the prospective designer
transformers and then finally filtered back to DC at its
an overview of the issues involved in designing
output. Switching power supplies are much more
switchmode power supplies. It describes the basic
efficient, ranging from 65 to 95 percent.
operation of the more popular topologies of switching
The downside of a switching design is that it is
power supplies, their relevant parameters, provides
considerably more complex. In addition, the output
circuit design tips, and information on how to select the
voltage contains switching noise, which must be
most appropriate semiconductor and passive
removed for many applications.
components. The guide also lists the ON Semiconductor
Although there are clear differences between linear
components expressly built for use in switching power
and switching regulators, many applications require both
supplies.
types to be used. For example, a switching regulator may
provide the initial regulation, then a linear regulator may
Linear versus Switching
provide post regulation for a noise sensitive part of the
Power Supplies
design, such as a sensor interface circuit.
Switching and linear regulators use fundamentally
different techniques to produce a regulated output
Switching Power Supply
voltage from an unregulated input. Each technique has
advantages and disadvantages, so the application will
Fundamentals
determine the most suitable choice.
There are two basic types of pulse width modulated
Linear power supplies can only step down an input
(PWM) switching power supplies, forward mode and
voltage to produce a lower output voltage. This is done
boost mode. They differ in the way the magnetic
by operating a bipolar transistor or MOSFET pass unit in
elements are operated. Each basic type has its advantages
its linear operating mode; that is, the drive to the pass unit
and disadvantages.
is proportionally changed to maintain the required output
voltage. Operating in this mode means that there is
The Forward Mode Converter
always a headroom voltage, Vdrop, between the input
The forward mode converter can be recognized by the
and the output. Consequently the regulator dissipates a
presence of an L C filter on its output. The L C filter
considerable amount of power, given by (Vdrop Iload).
creates a DC output voltage, which is essentially the
This headroom loss causes the linear regulator to only
volt time average of the L C filter s input AC
be 35 to 65 percent efficient. For example, if a 5.0 V
rectangular waveform. This can be expressed as:
regulator has a 12 V input and is supplying 100 mA, it
(eq. 1)
Vout [ Vin @ duty cycle
must dissipate 700 mW in the regulator in order to deliver
500 mW to the load , an efficiency of only 42 percent. The switching power supply controller varies the duty
The cost of the heatsink actually makes the linear cycle of the input rectangular voltage waveform and thus
regulator uneconomical above 10 watts for small controls the signal s volt time average.
applications. Below that point, however, linear The buck or step down converter is the simplest
regulators are cost effective in step down applications. forward mode converter, which is shown in Figure 1.
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SMPSRM
LO
SW
Vin Ion D Ioff Rload
Cout
Vsat
Power Power
Switch Switch
Power Power
OFF OFF
Switch Switch
ON ON
TIME
Vfwd
Ipk
Iload
Imin
Power SW Diode Power SW Diode
TIME
Figure 1. A Basic Forward Mode Converter and Waveforms (Buck Converter Shown)
Its operation can be better understood when it is broken clamped when the catch diode D becomes forward
into two time periods: when the power switch is turned biased. The stored energy then continues flowing to the
on and turned off. When the power switch is turned on, output through the catch diode and the inductor. The
the input voltage is directly connected to the input of the inductor current decreases from an initial value ipk and is
L C filter. Assuming that the converter is in a given by:
steady state, there is the output voltage on the filter s
Voutt
iL(off) + ipk * 0 v t v toff
(eq. 3)
output. The inductor current begins a linear ramp from an
L
initial current dictated by the remaining flux in the
The off period continues until the controller turns the
inductor. The inductor current is given by:
power switch back on and the cycle repeats itself.
( )
Vin * Vout
The buck converter is capable of over one kilowatt of
iL(on) + t ) iinit 0 v t v ton
(eq. 2)
L
output power, but is typically used for on board regulator
During this period, energy is stored as magnetic flux applications whose output powers are less than 100 watts.
within the core of the inductor. When the power switch Compared to the flyback mode converter, the forward
is turned off, the core contains enough energy to supply converter exhibits lower output peak to peak ripple
the load during the following off period plus some voltage. The disadvantage is that it is a step down
reserve energy. topology only. Since it is not an isolated topology, for
When the power switch turns off, the voltage on the safety reasons the forward converter cannot be used for
input side of the inductor tries to fly below ground, but is input voltages greater than 42.5 VDC.
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(VOLTS)
DIODE VOLTAGE
(AMPS)
INDUCTOR CURRENT
SMPSRM
The Flyback Mode Converter different fashion from the forward mode converter. The
most elementary flyback mode converter, the boost or
The basic flyback mode converter uses the same
step up converter, is shown in Figure 2.
components as the basic forward mode converter, but in
a different configuration. Consequently, it operates in a
L
D
Cout
Vin Rload
SW
Iload
Ion Ioff
Vin
Vflbk
Power Power
(Vout)
Switch Switch
ON ON
Power
Vsat Diode Switch Diode
ON ON ON
TIME
Ipk
Iload
TIME
Figure 2. A Basic Boost Mode Converter and Waveforms (Boost Converter Shown)
Again, its operation is best understood by considering the the output rectifier when its voltage exceeds the output
on and off periods separately. When the power voltage. The energy within the core of the inductor is then
switch is turned on, the inductor is connected directly passed to the output capacitor. The inductor current
across the input voltage source. The inductor current then during the off period has a negative ramp whose slope is
rises from zero and is given by: given by:
Vint
(Vin * Vout)
iL(on) + v t v 0on (eq. 4)
L
iL(off) +
(eq. 6)
L
Energy is stored within the flux in the core of the inductor.
The peak current, ipk, occurs at the instant the power
The energy is then completely emptied into the output
switch is turned off and is given by:
capacitor and the switched terminal of the inductor falls
Vin ton back to the level of the input voltage. Some ringing is
ipk +
(eq. 5)
L evident during this time due to residual energy flowing
When the power switch turns off, the switched side of through parasitic elements such as the stray inductances
the inductor wants to fly up in voltage, but is clamped by and capacitances in the circuit.
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(VOLTS)
SWITCH VOLTAGE
(AMPS)
INDUCTOR CURRENT
SMPSRM
When there is some residual energy permitted to to a 50 percent duty cycle. There must be a time period
remain within the inductor core, the operation is called when the inductor is permitted to empty itself of its
continuous mode. This can be seen in Figure 3. energy.
The boost converter is used for board level (i.e.,
Energy for the entire on and off time periods must be
non isolated) step up applications and is limited to less
stored within the inductor. The stored energy is defined
than 100 150 watts due to high peak currents. Being a
by:
non isolated converter, it is limited to input voltages of
(eq. 7)
EL + 0.5L @ ipk2
less than 42.5 VDC. Replacing the inductor with a
The boost mode inductor must store enough energy to transformer results in a flyback converter, which may be
supply the output load for the entire switching period (ton step up or step down. The transformer also provides
+ toff). Also, boost mode converters are typically limited dielectric isolation from input to output.
Vflbk
(Vout)
Vin
Power Power
Switch Diode Switch Diode
ON ON ON ON
TIME
Vsat
Ipk
TIME
Figure 3. Waveforms for a Continuous Mode Boost Converter
5. How much of the input voltage is placed across
Common Switching
the primary transformer winding or inductor?
Power Supply Topologies
Factor 1 is a safety related issue. Input voltages above
A topology is the arrangement of the power devices
42.5 VDC are considered hazardous by the safety
and their magnetic elements. Each topology has its own
regulatory agencies throughout the world. Therefore,
merits within certain applications. There are five major
only transformer isolated topologies must be used above
factors to consider when selecting a topology for a
this voltage. These are the off line applications where the
particular application. These are:
power supply is plugged into an AC source such as a wall
1. Is input to output dielectric isolation required for
socket.
the application? This is typically dictated by the
Multiple outputs require a transformer based
safety regulatory bodies in effect in the region.
topology. The input and output grounds may be
2. Are multiple outputs required?
connected together if the input voltage is below
3. Does the prospective topology place a reasonable
42.5 VDC. Otherwise full dielectric isolation is required.
voltage stress across the power semiconductors?
4. Does the prospective topology place a reasonable
current stress upon the power semiconductors?
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(VOLTS)
SWITCH VOLTAGE
(AMPS)
INDUCTOR CURRENT
SMPSRM
Factors 3, 4 and 5 have a direct affect upon the Cost is a major factor that enters into the topology
reliability of the system. Switching power supplies decision. There are large overlaps in the performance
deliver constant power to the output load. This power is boundaries between the topologies. Sometimes the most
then reflected back to the input, so at low input voltages, cost effective choice is to purposely design one topology
the input current must be high to maintain the output to operate in a region that usually is performed by
power. Conversely, the higher the input voltage, the another. This, though, may affect the reliability of the
lower the input current. The design goal is to place as desired topology.
much as possible of the input voltage across the Figure 4 shows where the common topologies are used
transformer or inductor so as to minimize the input for a given level of DC input voltage and required output
current. power. Figures 5 through 12 show the common
Boost mode topologies have peak currents that are topologies. There are more topologies than shown, such
about twice those found in forward mode topologies. as the Sepic and the Cuk, but they are not commonly
This makes them unusable at output powers greater than used.
100 150 watts.
1000
Half Bridge
Flyback
Full Bridge
100
42.5
Full Bridge
Non Isolated
10
Very High
Buck
Peak Currents
10 100 1000
OUTPUT POWER (W)
Figure 4. Where Various Topologies Are Used
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DC INPUT VOLTAGE (V)
SMPSRM
L
VFWD
Power Switch VD
+
+
0 TIME
Vin
D
IPK
Vout
+
Vin Cin
Cout
Control
IL
Feedback
0 TIME
ILOAD IMIN
Figure 5. The Buck (Step Down) Converter
VFLBK
D D
VSAT ON ON
VSW
L
SW ON
D
Vin Cin TIME
0
+
Vin
+
SW
Control
Vout
Cout
IPK
IL
ISW ID
TIME
0
Figure 6. The Boost (Step Up) Converter
+
Vin
VL 0
TIME
Control
SW
D
Vin
Cin
Vout
+
L Vout
Cout
+ IL
Feedback
ISW ID
0 TIME
IPK
Figure 7. The Buck Boost (Inverting) Converter
VFLBK
VSAT
SW
ON
VSW
TIME
0
Vin
+ D
+
+
N1 N2
Cout Vout IPRI
Cin
Vin 0 TIME
IPK
SW
Control
ISEC
Feedback
0 TIME
Figure 8. The Flyback Converter
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SMPSRM
LO
+
D
T
+
+
N2
N1
Cout Vout
Cin
Vin
SW
Control
Feedback
SW
ON
VSW
TIME
0
VSAT 2Vin
IPRI
0 TIME
IMIN IPK
Figure 9. The One Transistor Forward Converter (Half Forward Converter)
LO
SW1
D1
T
+
+
+
Cout Vout
D2
SW2
Vin
Cin Control
Feedback
2Vin
Vin SW2
VSW
SW1
TIME
0
VSAT
IPK
IPRI
0
TIME
IMIN
Figure 10. The Push Pull Converter
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SMPSRM
LO
Ds +
+
Cout
Vout
+
N2
XFMR
SW1
C
T
Cin
Control
N1
Vin
C
SW2
Feedback
Vin
SW1
Vin
2
SW2
VSW2 0
TIME
VSAT
IPK
IPRI
TIME
0
IMIN
Figure 11. The Half Bridge Converter
LO
Ds +
+
Cout
Vout
+
N2
XFMR XFMR
SW1 SW3
T
Cin
N1
Vin
Control
C
SW2 SW4
Vin
SW
Vin
1-4
2
SW
2-3
VSW2 0
TIME
VSAT
IPK
ISW2
0
TIME
IMIN
Figure 12. The Full Bridge Converter
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SMPSRM
The input and output capacitors are shared among the
Interleaved Multiphase Converters
phases. The input capacitor sees less RMS ripple current
One method of increasing the output power of any
because the peak currents are less and the combined duty
topology and reducing the stresses upon the
cycle of the phases is greater than it would experience
semiconductors, is a technique called interleaving. Any
with a single phase converter. The output capacitor can
topology can be interleaved. An interleaved multiphase
be made smaller because the frequency of current
converter has two or more identical converters placed in
waveform is n times higher and its combined duty cycle
parallel which share key components. For an n phase
is greater. The semiconductors also see less current
converter, each converter is driven at a phase difference
stress.
of 360/n degrees from the next. The output current from
A block diagram of an interleaved multiphase buck
all the phases sum together at the output, requiring only
converter is shown in Figure 13. This is a 2 phase
Iout/n amperes from each phase.
topology that is useful in providing power to a high
performance microprocessor.
+
+
VIN
CIN
SA1
LA
SA2
VFDBK
GATEA1
Control GATEA2 +
+
COUT VOUT
GND GATEB2
SB1
CFA GATEB1 LB
CFB
CS5308
SB2
Current Feedback A
Current Feedback B
Voltage Feedback
Figure 13. Example of a Two Phase Buck Converter with Voltage and Current Feedback
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select the one that is wanted.
Selecting the Method of Control
Table 1 summarizes the features of each of the popular
There are three major methods of controlling a
methods of control. Certain methods are better adapted to
switching power supply. There are also variations of
certain topologies due to reasons of stability or transient
these control methods that provide additional protection
response.
features. One should review these methods carefully and
then carefully review the controller IC data sheets to
Table 1. Common Control Methods Used in ICs
Control Method OC Protection Response Time Preferred Topologies
Average OC Slow Forward Mode
Voltage Mode
Voltage Mode
Pulse by Pulse OC Slow Forward Mode
Intrinsic Rapid Boost Mode
Current Mode
Current Mode
Hysteretic Rapid Boost & Forward Mode
Hysteric Voltage Average Slow Boost & Forward Mode
Voltage mode control (see Figure 14) is typically used instantly cutoff if its limits are exceeded. This offers
better protection to the power switch.
for forward mode topologies. In voltage mode control,
Current mode control (see Figure 15) is typically used
only the output voltage is monitored. A voltage error
with boost mode converters. Current mode control
signal is calculated by forming the difference between
monitors not only the output voltage, but also the output
Vout (actual) and Vout(desired). This error signal is then
current. Here the voltage error signal is used to control
fed into a comparator that compares it to the ramp voltage
the peak current within the magnetic elements during
generated by the internal oscillator section of the control
each power switch on time. Current mode control has a
IC. The comparator thus converts the voltage error signal
very rapid input and output response time, and has an
into the PWM drive signal to the power switch. Since the
inherent overcurrent protection. It is not commonly used
only control parameter is the output voltage, and there is
for forward mode converters; their current waveforms
inherent delay through the power circuit, voltage mode
have much lower slopes in their current waveforms
control tends to respond slowly to input variations.
which can create jitter within comparators.
Overcurrent protection for a voltage mode controlled
Hysteretic control is a method of control which tries to
converter can either be based on the average output
keep a monitored parameter between two limits. There
current or use a pulse by pulse method. In average
are hysteretic current and voltage control methods, but
overcurrent protection, the DC output current is
they are not commonly used.
monitored, and if a threshold is exceeded, the pulse width
The designer should be very careful when reviewing a
of the power switch is reduced. In pulse by pulse
prospective control IC data sheet. The method of control
overcurrent protection, the peak current of each power
and any variations are usually not clearly described on
switch on cycle is monitored and the power switch is the first page of the data sheet.
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VCC
OSC Charge
Clock Ramp
Verror
Discharge
Ct
Volt
Output
Verror Amp.
Comp.
Gating
Logic
VFB
+
+
+
Pulsewidth
Vref
Comparator
Steering
Average
Overcurrent
Protection
Cur.
Comp.
Current Amp.
Iout (lavOC) +
Pulse by Pulse
or RCS
+
Overcurrent
ISW (P POC) VOC
Protection
VSS
Figure 14. Voltage Mode Control
VCC
OSC
+
Ct
Discharge
Output
Gating
Logic
Volt
S
Verror Amp.
Output
Comp. Q
VFB
Verror R
+
+
Vref
Current
S R S
Comparator
+
Verror
ISW
RCS
Ipk
VSS
ISW
Figure 15. Turn On with Clock Current Mode Control
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One should generate a gate drive voltage that is as close
The Choice of Semiconductors
to 0.7 volts as possible. This is to minimize any loss
Power Switches
created by dropping the base drive voltage at the required
The choice of which semiconductor technology to use
base current to the level exhibited by the base.
for the power switch function is influenced by many
A second consideration is the storage time exhibited by
factors such as cost, peak voltage and current, frequency
the collector during its turn off transition. When the base
of operation, and heatsinking. Each technology has its
is overdriven, or where the base current is more than
own peculiarities that must be addressed during the
needed to sustain the collector current, the collector
design phase.
exhibits a 0.3 2 ms delay in its turn off which is
There are three major power switch choices: the
proportional to the base overdrive. Although the storage
bipolar junction transistor (BJT), the power MOSFET,
time is not a major source of loss, it does significantly
and the integrated gate bipolar transistor (IGBT). The
limit the maximum switching frequency of a
BJT was the first power switch to be used in this field and
still offers many cost advantages over the others. It is also bipolar based switching power supply. There are two
still used for very low cost or in high power switching methods of reducing the storage time and increasing its
converters. The maximum frequency of operation of switching time. The first is to use a base speed up
bipolar transistors is less than 80 100 kHz because of capacitor whose value, typically around 100 pF, is placed
some of their switching characteristics. The IGBT is used in parallel with the base current limiting resistor
for high power switching converters, displacing many of (Figure 16a). The second is to use proportional base drive
the BJT applications. They too, though, have a slower (Figure 16b). Here, only the amount of needed base
switching characteristic which limits their frequency of current is provided by the drive circuit by bleeding the
operation to below 30 kHz typically although some can
excess around the base into the collector.
reach 100 kHz. IGBTs have smaller die areas than power
The last consideration with BJTs is the risk of
MOSFETs of the same ratings, which typically means a
excessive second breakdown. This phenomenon is
lower cost. Power MOSFETs are used in the majority of
caused by the resistance of the base across the die,
applications due to their ease of use and their higher
permitting the furthest portions of the collector to turn off
frequency capabilities. Each of the technologies will be
later. This forces the current being forced through the
reviewed.
collector by an inductive load, to concentrate at the
opposite ends of the die, thus causing an excessive
The Bipolar Power Transistor
localized heating on the die. This can result in a
The BJT is a current driven device. That means that the
short circuit failure of the BJT which can happen
base current is in proportion to the current drawn through
instantaneously if the amount of current crowding is
the collector. So one must provide:
great, or it can happen later if the amount of heating is
IB u IC hFE (eq. 8)
less. Current crowding is always present when an
In power transistors, the average gain (hFE) exhibited at inductive load is attached to the collector. By switching
the higher collector currents is between 5 and 20. This the BJT faster, with the circuits in Figure 15, one can
could create a large base drive loss if the base drive circuit greatly reduce the effects of second breakdown on the
is not properly designed. reliability of the device.
VBB
VBB
+
100 pF
VCE
Control IC
+
VBE
100 pF
Control IC
Power Ground
Power Ground
(a) Fixed Base Drive Circuit (b) Proportional Base Drive Circuit (Baker Clamp)
Figure 16. Driving a Bipolar Junction Transistor
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From the gate terminal, there are two capacitances the
The Power MOSFET
designer encounters, the gate input capacitance (Ciss) and
Power MOSFETs are the popular choices used as
the drain gate reverse capacitance (Crss). The gate input
power switches and synchronous rectifiers. They are, on
capacitance is a fixed value caused by the capacitance
the surface, simpler to use than BJTs, but they have some
formed between the gate metalization and the substrate.
hidden complexities.
Its value usually falls in the range of 800 3200 pF,
A simplified model for a MOSFET can be seen in
depending upon the physical construction of the
Figure 17. The capacitances seen in the model are
MOSFET. The Crss is the capacitance between the drain
specified within the MOSFET data sheets, but can be
and the gate, and has values in the range of 60 150 pF.
nonlinear and vary with their applied voltages.
Although the Crss is smaller, it has a much more
pronounced effect upon the gate drive. It couples the
drain voltage to the gate, thus dumping its stored charge
into the gate input capacitance. The typical gate drive
waveforms can be seen in Figure 18. Time period t1 is
only the Ciss being charged or discharged by the
CDG
impedance of the external gate drive circuit. Period t2
Coss
shows the effect of the changing drain voltage being
coupled into the gate through Crss. One can readily
CGS
observe the flattening of the gate drive voltage during
this period, both during the turn on and turn off of the
MOSFET. Time period t3 is the amount of overdrive
voltage provided by the drive circuit but not really
Figure 17. The MOSFET Model
needed by the MOSFET.
VDR
TURN TURN OFF
ON
t3 t3
t1 t2 t2 t1
VGS
Vpl
Vth
0
VDS
0
IG +
0
Figure 18. Typical MOSFET Drive Waveforms (Top: VGS, Middle: VDG, Bottom: IG)
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The time needed to switch the MOSFET between on Driving MOSFETs in Switching
and off states is dependent upon the impedance of the Power Supply Applications
gate drive circuit. It is very important that the drive circuit There are three things that are very important in the
be bypassed with a capacitor that will keep the drive high frequency driving of MOSFETs: there must be a
voltage constant over the drive period. A 0.1 mF capacitor totem pole driver; the drive voltage source must be well
is more than sufficient. bypassed; and the drive devices must be able to source
high levels of current in very short periods of time (low
compliance). The optimal drive circuit is shown in
Figure 19.
VG VG
LOAD LOAD
Ron
Roff
a. Passive Turn ON b. Passive Turn OFF
VG VG
LOAD LOAD
c. Bipolar Totem pole d. MOS Totem pole
Figure 19. Bipolar and FET Based Drive Circuits (a. Bipolar Drivers, b. MOSFET Drivers)
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Sometimes it is necessary to provide a circuit. Both of the series capacitors must be more than
dielectrically isolated drive to a MOSFET. This is 10 times the value of the Ciss of the MOSFET so that the
provided by a drive transformer. Transformers driven capacitive voltage divider that is formed by the series
from a DC source must be capacitively coupled from the capacitors does not cause an excessive attenuation. The
totem pole driver circuit. The secondary winding must circuit can be seen in Figure 20.
be capacitively coupled to the gate with a DC restoration
C
RG
T
VG
1 k
C
1:1
C > 10
Ciss
Figure 20. Transformer Isolated Gate Drive
Rectifiers
The Insulated Gate Bipolar
Rectifiers represent about 60 percent of the losses in
Transistor (IGBT)
nonsynchronous switching power supplies. Their choice
The IGBT is a hybrid device with a MOSFET as the
has a very large effect on the efficiency of the power
input device, which then drives a silicon controlled
supply.
rectifier (SCR) as a switched output device. The SCR is
The significant rectifier parameters that affect the
constructed such that it does not exhibit the latching
operation of switching power supplies are:
characteristic of a typical SCR by making its feedback
" forward voltage drop (Vf), which is the voltage
gain less than 1. The die area of the typical IGBT is less
across the diode when a forward current is flowing
than one half that of an identically rated power MOSFET,
" the reverse recovery time (trr), which is how long it
which makes it less expensive for high power converters.
requires a diode to clear the minority charges from
The only drawback is the turn off characteristic of the
its junction area and turn off when a reverse voltage
IGBT. Being a bipolar minority carrier device, charges
must be removed from the P N junctions during a turn off is applied
condition. This causes a current tail at the end of the
" the forward recovery time (tfrr) which is how long it
turn off transition of the current waveform. This can be a
take a diode to begin to conduct forward current
significant loss because the voltage across the IGBT is
after a forward voltage is applied.
very high at that moment. This makes the IGBT useful
There are four choices of rectifier technologies:
only for frequencies typically less than 20 kHz, or for
standard, fast and ultra fast recovery types, and Schottky
exceptional IGBTs, 100 kHz.
barrier types.
To drive an IGBT one uses the MOSFET drive circuits
A standard recovery diode is only suitable for
shown in Figures 18 and 19. Driving the IGBT gate faster
50 60 Hz rectification due to its slow turn off
makes very little difference in the performance of an
characteristics. These include common families such as
IGBT, so some reduction in drive currents can be used.
the 1N4000 series diodes. Fast recovery diodes were
The voltage drop of across the collector to emitter
first used in switching power supplies, but their turn off
(VCE) terminals is comparable to those found in
time is considered too slow for most modern
Darlington BJTs and MOSFETs operated at high currents.
applications. They may find application where low cost
The typical VCE of an IGBT is a flat 1.5 2.2 volts.
is paramount, however. Ultra fast recovery diodes turn
MOSFETs, acting more resistive, can have voltage drops off quickly and have a forward voltage drop of 0.8 to
of up to 5 volts at the end of some high current ramps. This 1.3 V, together with a high reverse voltage capability of
makes the IGBT, in high current environments, very up to 1000 V. A Schottky rectifier turns off very quickly
comparable to MOSFETs in applications of less than and has an average forward voltage drop of between 0.35
5 30 kHz. and 0.8 V, but has a low reverse breakdown voltage and
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a high reverse leakage current. For a typical switching The characteristics of power rectifiers and their
power supply application, the best choice is usually a applications in switching power supplies are covered in
Schottky rectifier for output voltages less than 12 V, and great detail in Reference (5).
an ultra fast recovery diode for all other output voltages. The major losses within output rectifiers are
The major losses within output rectifiers are conduction losses and switching losses. The conduction
conduction losses and switching losses. The conduction loss is the forward voltage drop times the current flowing
loss is the forward voltage drop times the current flowing through it during its conduction period. This can be
through it during its conduction period. This can be significant if its voltage drop and current are high. The
significant if its voltage drop and current are high. The switching losses are determined by how fast a diode turns
switching losses are determined by how fast a diode turns off (trr) times the reverse voltage across the rectifier. This
off (trr) times the reverse voltage across the rectifier. This can be significant for high output voltages and currents.
can be significant for high output voltages and currents.
Table 2. Types of Rectifier Technologies
Rectifier Type Average Vf Reverse Recovery Time Typical Applications
Standard Recovery 0.7 1.0 V 1,000 ns 50 60 Hz Rectification
Fast Recovery 1.0 1.2 V 150 200 ns Output Rectification
Output Rectification
UltraFast Recovery 0.9 1.4 V 25 75 ns
(Vo > 12 V)
Output Rectification
Schottky 0.3 0.8 V < 10 ns
(Vo < 12 V)
Table 3. Estimating the Significant Parameters of the Power Semiconductors
Bipolar Pwr Sw MOSFET Pwr Sw Rectifier
Topology
Topology
VCEO IC VDSS ID VR IF
Buck Vin Iout Vin Iout Vin Iout
(2.0 Pout) (2.0 Pout)
Boost Vout Vout Vout Iout
Vin(min) Vin(min)
(2.0 Pout)
( )
2.0 Pout
Buck/Boost
Vin * Vout Vin(min) Vin * Vout Vin(min) Vin * Vout Iout
(2.0 Pout) (2.0 Pout)
Flyback 5.0 Vout Iout
1.7 Vin(max) Vin(min) 1.5 Vin(max) Vin(min)
1 Transistor (1.5 Pout) (1.5 Pout)
2.0 Vin 2.0 Vin 3.0 Vout Iout
Forward
Vin(min) Vin(min)
(1.2 Pout)
(1.2 Pout)
Push Pull 2.0 Vin 2.0 Vin 2.0 Vout Iout
Vin(min)
Vin(min)
(2.0 Pout) (2.0 Pout)
Half Bridge Vin Vin 2.0 Vout Iout
Vin(min) Vin(min)
(1.2 Pout) (2.0 Pout)
Full Bridge Vin Vin 2.0 Vout Iout
Vin(min) Vin(min)
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Coiltronics, Division of Cooper Electronics
The Magnetic Components
Technology
The magnetic elements within a switching power
6000 Park of Commerce Blvd
supply are used either for stepping up or down a
Boca Raton, FL (USA) 33487
switched AC voltage, or for energy storage. In
website: http://www.coiltronics.com
forward mode topologies, the transformer is only used
Telephone: 561 241 7876
for stepping up or down the AC voltage generated by the
power switches. The output filter (the output inductor
Cramer Coil, Inc.
and capacitor) in forward mode topologies is used for
401 Progress Dr.
energy storage. In boost mode topologies, the
Saukville, WI (USA) 53080
transformer is used both for energy storage and to provide
website: http://www.cramerco.com
a step up or step down function.
email: techsales@cramercoil.com
Many design engineers consider the magnetic
Telephone: 262 268 2150
elements of switching power supplies counter intuitive
or too complicated to design. Fortunately, help is at hand;
Pulse, Inc.
the suppliers of magnetic components have applications
San Diego, CA
engineers who are quite capable of performing the
website: http://www.pulseeng.com
transformer design and discussing the tradeoffs needed
Telephone: 858 674 8100
for success. For those who are more experienced or more
TDK
adventuresome, please refer to Reference 2 in the
1600 Feehanville Drive
Bibliography for transformer design guidelines.
Mount Prospect, IL 60056
The general procedure in the design of any magnetic
website: http://www.component.talk.com
component is as follows (Reference 2, p 42):
Telephone: 847 803 6100
1. Select an appropriate core material for the
application and the frequency of operation.
2. Select a core form factor that is appropriate for
the application and that satisfies applicable
regulatory requirements.
Laying Out the Printed Circuit Board
3. Determine the core cross sectional area
The printed circuit board (PCB) layout is the third
necessary to handle the required power
critical portion of every switching power supply design
4. Determine whether an airgap is needed and
in addition to the basic design and the magnetics design.
calculate the number of turns needed for each
Improper layout can adversely affect RFI radiation,
winding. Then determine whether the accuracy
component reliability, efficiency and stability. Every
of the output voltages meets the requirements
PCB layout will be different, but if the designer
and whether the windings will fit into the
appreciates the common factors present in all switching
selected core size.
power supplies, the process will be simplified.
5. Wind the magnetic component using proper
All PCB traces exhibit inductance and resistance.
winding techniques.
These can cause high voltage transitions whenever there
6. During the prototype stage, verify the
is a high rate of change in current flowing through the
component s operation with respect to the level
trace. For operational amplifiers sharing a trace with
of voltage spikes, cross regulation, output
power signals, it means that the supply would be
accuracy and ripple, RFI, etc., and make
impossible to stabilize. For traces that are too narrow for
corrections were necessary.
the current flowing through them, it means a voltage drop
The design of any magnetic component is a calculated
from one end of the trace to the other which potentially
estimate. There are methods of stretching the design
can be an antenna for RFI. In addition, capacitive
limits for smaller size or lower losses, but these tend to
coupling between adjacent traces can interfere with
be diametrically opposed to one another. One should be
proper circuit operation.
cautious when doing this.
There are two rules of thumb for PCB layouts: short
Some useful sources for magnetics components are:
and fat for all power carrying traces and one point
CoilCraft, Inc. grounding for the various ground systems within a
1102 Silver Lake Rd. switching power supply. Traces that are short and fat
Cary, IL (USA) 60013 minimize the inductive and resistive aspects of the trace,
website: http://www.coilcraft.com/ thus reducing noise within the circuits and RFI.
email: info@coilcraft.com Single point grounding keeps the noise sources
Telephone: 847 639 6400 separated from the sensitive control circuits.
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Within all switching power supplies, there are four rectifier to the output filter capacitor and back to the
major current loops. Two of the loops conduct the inductor or winding. The filter capacitors are the only
high level AC currents needed by the supply. These are components that can source and sink the large levels of
the power switch AC current loop and the output rectifier AC current in the time needed by the switching power
AC current loop. The currents are the typical trapezoidal supply. The PCB traces should be made as wide and as
current pulses with very high peak currents and very short as possible, to minimize resistive and inductive
rapid di/dts. The other two current loops are the input effects. These traces should be the first to be laid out.
source and the output load current loops, which carry low
Turning to the input source and output load current
frequency current being supplied from the voltage source
loops, both of these loops must be connected directly to
and to the load respectively.
their respective filter capacitor s terminals, otherwise
For the power switch AC current loop, current flows switching noise could bypass the filtering action of the
from the input filter capacitor through the inductor or capacitor and escape into the environment. This noise is
transformer winding, through the power switch and back called conducted interference. These loops can be seen
to the negative pin of the input capacitor. Similarly, the in Figure 21 for the two major forms of switching
output rectifier current loop s current flows from the power supplies, non isolated (Figure 21a) and
inductor or secondary transformer winding, through the transformer isolated (Figure 21b).
Power Switch Output Rectifier
Current Loop L Current Loop
SW Vout
Input Current
Loop
Output Load
Current Loop
+
VFB
Control
Vin
Cin Cout
Analog GND
C
AB
Input Source Output Load
Power Output Rectifier
Ground Ground
Switch Ground Ground
Join Join
Join
(a) The Non Isolated DC/DC Converter
Output Rectifier Output Load
Input Current Power Switch
Current Loop Current Loop
Loop Current Loop
Vout
VFB
Cout
SW
+
Vin
Cin
Control
B
Output Rectifier Output Load
Ground Ground
RCS
Analog
Join
GND
FB
Join
C
A
Input Source
Power Switch Ground
Ground
Join (b) The Transformer Isolated Converter
Figure 21. The Current Loops and Grounds for the Major Converter Topologies
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The grounds are extremely important to the proper The last important factor in the PCB design is the
operation of the switching power supply, since they form layout surrounding the AC voltage nodes. These are the
the reference connections for the entire supply; each
drain of the power MOSFET (or collector of a BJT) and
ground has its own unique set of signals which can
the anode of the output rectifier(s). These nodes can
adversely affect the operation of the supply if connected
capacitively couple into any trace on different layers of
improperly.
the PCB that run underneath the AC pad. In surface
There are five distinct grounds within the typical
mount designs, these nodes also need to be large enough
switching power supply. Four of them form the return
to provide heatsinking for the power switch or rectifier.
paths for the current loops described above. The
This is at odds with the desire to keep the pad as small as
remaining ground is the low level analog control ground
possible to discourage capacitive coupling to other
which is critical for the proper operation of the supply.
traces. One good compromise is to make all layers below
The grounds which are part of the major current loops
the AC node identical to the AC node and connect them
must be connected together exactly as shown in
with many vias (plated through holes). This greatly
Figure 21. Here again, the connecting point between the
increases the thermal mass of the pad for improved
high level AC grounds and the input or output grounds
heatsinking and locates any surrounding traces off
is at the negative terminal of the appropriate filter
laterally where the coupling capacitance is much smaller.
capacitor (points A and B in Figures 21a and 21b). Noise
An example of this can be seen in Figure 22.
on the AC grounds can very easily escape into the
Many times it is necessary to parallel filter capacitors
environment if the grounds are not directly connected to
to reduce the amount of RMS ripple current each
the negative terminal of the filter capacitor(s). The
capacitor experiences. Close attention should be paid to
analog control ground must be connected to the point
this layout. If the paralleled capacitors are in a line, the
where the control IC and associated circuitry must
capacitor closest to the source of the ripple current will
measure key power parameters, such as AC or DC
operate hotter than the others, shortening its operating
current and the output voltage (point C in Figures 21a and
life; the others will not see this level of AC current. To
21b). Here any noise introduced by large AC signals
ensure that they will evenly share the ripple current,
within the AC grounds will sum directly onto the
ideally, any paralleled capacitors should be laid out in a
low level control parameters and greatly affect the
radially symmetric manner around the current source,
operation of the supply. The purpose of connecting the
typically a rectifier or power switch.
control ground to the lower side of the current sensing
The PCB layout, if not done properly, can ruin a good
resistor or the output voltage resistor divider is to form a
paper design. It is important to follow these basic
Kelvin contact where any common mode noise is not
sensed by the control circuit. In short, follow the example guidelines and monitor the layout every step of the
given by Figure 21 exactly as shown for best results. process.
Power Device
Via PCB Top
Plated Thru Hole PCB Bottom
Figure 22. Method for Minimizing AC Capacitive Coupling and Enhancing Heatsinking
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the circuitry, and some are controlled by simply selecting
Losses and Stresses in Switching
a different part. Identifying the major sources for loss can
Power Supplies
be as easy as placing a finger on each of the components
Much of the designer s time during a switching power
in search of heat, or measuring the currents and voltages
supply design is spent in identifying and minimizing the
associated with each power component using an
losses within the supply. Most of the losses occur in the
oscilloscope, AC current probe and voltage probe.
power components within the switching power supply.
Semiconductor losses fall into two categories:
Some of these losses can also present stresses to the
conduction losses and switching losses. The conduction
power semiconductors which may affect the long term
loss is the product of the terminal voltage and current
reliability of the power supply, so knowing where they
during the power device s on period. Examples of
arise and how to control them is important.
conduction losses are the saturation voltage of a bipolar
Whenever there is a simultaneous voltage drop across
a component with a current flowing through, there is a power transistor and the on loss of a power MOSFET
loss. Some of these losses are controllable by modifying shown in Figure 23 and Figure 24 respectively.
VPEAK
VPEAK
SATURATION
FALL
VOLTAGE
ON VOLTAGE
TIME
RISE DYNAMIC STORAGE
RISE FALL
TIME SATURATION TIME
TIME TIME
CLEARING
IPEAK
CLEARING
RECTIFIERS
RECTIFIERS IPEAK
PINCHING OFF INDUCTIVE
PINCHING OFF INDUCTIVE
CHARACTERISTICS OF THE
CHARACTERISTICS OF THE
TRANSFORMER
TRANSFORMER
ON CURRENT
SATURATION
CURRENT
TURN-ON TURN-OFF CURRENT TURN-ON TURN-OFF
CURRENT CURRENT TAIL CURRENT CURRENT
CURRENT
CROWDING
SECOND
PERIOD
BREAKDOWN
PERIOD
ON LOSS
SATURATION
LOSS
TURN-ON TURN-OFF LOSS
TURN-ON TURN-OFF LOSS
LOSS SWITCHING LOSS
LOSS SWITCHING LOSS
Figure 23. Stresses and Losses Figure 24. Stresses and Losses
within a Bipolar Power Transistor within a Power MOSFET
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(VOLTS)
(VOLTS)
COLLECTOR-TO-EMITTER
DRAIN-TO-SOURCE VOLTAGE
(AMPS)
(AMPS)
DRAIN CURRENT
COLLECTOR CURRENT
LOSS (JOULES)
LOSS (JOULES)
INSTANTANEOUS ENERGY
INSTANTANEOUS ENERGY
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The forward conduction loss of a rectifier is shown in creates a very large V I product which is as significant as
Figure 25. During turn off, the rectifier exhibits a reverse the conduction losses. Switching losses are also the major
recovery loss where minority carriers trapped within the frequency dependent loss within every PWM switching
P N junction must reverse their direction and exit the power supply.
junction after a reverse voltage is applied. This results in The loss induced heat generation causes stress within
what appears to be a current flowing in reverse through the power component. This can be minimized by an
the diode with a high reverse terminal voltage. effective thermal design. For bipolar power transistors,
The switching loss is the instantaneous product of the however, excessive switching losses can also provide a
terminal voltage and current of a power device when it is lethal stress to the transistor in the form of second
transitioning between operating states (on to off and breakdown and current crowding failures. Care should be
off to on). Here, voltages are transitional between taken in the careful analysis of each transistor s Forward
full on and cutoff states while simultaneously the current Biased Safe Operating Area (FBSOA) and Reverse
is transitional between full on and cut off states. This Biased Safe Operating Area (RBSOA) operation.
FORWARD VOLTAGE
REVERSE VOLTAGE
IPK
FORWARD CONDUCTION CURRENT
DEGREE OF DIODE
RECOVERY
ABRUPTNESS
FORWARD REVERSE
RECOVERY RECOVERY
TIME (Tfr) TIME (Trr)
FORWARD CONDUCTION LOSS
SWITCHING
LOSS
Figure 25. Stresses and Losses within Rectifiers
rectification is a technique to reduce this conduction loss
Techniques to Improve Efficiency in
by using a switch in place of the diode. The synchronous
Switching Power Supplies
rectifier switch is open when the power switch is closed,
The reduction of losses is important to the efficient
and closed when the power switch is open, and is
operation of a switching power supply, and a great deal
typically a MOSFET inserted in place of the output
of time is spent during the design phase to minimize these
rectifier. To prevent crowbar current that would flow if
losses. Some common techniques are described below.
both switches were closed at the same time, the switching
scheme must be break before make. Because of this, a
The Synchronous Rectifier
diode is still required to conduct the initial current during
As output voltages decrease, the losses due to the
output rectifier become increasingly significant. For the interval between the opening of the main switch and
Vout = 3.3 V, a typical Schottky diode forward voltage of the closing of the synchronous rectifier switch. A
0.4 V leads to a 12% loss of efficiency. Synchronous Schottky rectifier with a current rating of 30 percent of
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25
(VOLTS)
DIODE VOLTAGE
(AMPS)
DIODE CURRENT
LOSS (JOULES)
INSTANTANEOUS ENERGY
SMPSRM
the MOSFET should be placed in parallel with the typical switching power supply.
synchronous MOSFET. The MOSFET does contain a The synchronous rectifier can be driven either actively,
parasitic body diode that could conduct current, but it is that is directly controlled from the control IC, or
lossy, slow to turn off, and can lower efficiency by 1% to passively, driven from other signals within the power
2%. The lower turn on voltage of the Schottky prevents circuit. It is very important to provide a non overlapping
the parasitic diode from ever conducting and exhibiting drive between the power switch(es) and the synchronous
its poor reverse recovery characteristic. rectifier(s) to prevent any shoot through currents. This
Using synchronous rectification, the conduction dead time is usually between 50 to 100 ns. Some typical
voltage can be reduced from 400 mV to 100 mV or less. circuits can be seen in Figure 26.
An improvement of 1 5 percent can be expected for the
Vin + Vout
SW
Drive
GND
Direct
SR
RG
C
VG
C
D
1 k
1:1
C > 10 Ciss
Transformer Isolated
(a) Actively Driven Synchronous Rectifiers
LO
+
Vout
Primary
(b) Passively Driven Synchronous Rectifiers
Figure 26. Synchronous Rectifier Circuits
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26
SMPSRM
Snubbers and Clamps Therefore it is not very useful for reducing RFI. It is
Snubbers and clamps are used for two very different useful for preventing components such as
purposes. When misapplied, the reliability of the semiconductors and capacitors from entering avalanche
semiconductors within the power supply is greatly breakdown.
jeopardized. Bipolar power transistors suffer from current crowding
A snubber is used to reduce the level of a voltage spike which is an instantaneous failure mode. If a voltage spike
and decrease the rate of change of a voltage waveform. occurs during the turn off voltage transition of greater
This then reduces the amount of overlap of the voltage than 75 percent of its VCEO rating, it may have too much
and current waveforms during a transition, thus reducing current crowding stress. Here both the rate of change of
the switching loss. This has its benefits in the Safe the voltage and the peak voltage of the spike must be
Operating Area (SOA) of the semiconductors, and it controlled. A snubber is needed to bring the transistor
reduces emissions by lowering the spectral content of any within its RBSOA (Reverse Bias Safe Operating Area)
RFI. rating. Typical snubber and clamp circuits are shown in
A clamp is used only for reducing the level of a voltage Figure 27. The effects that these have on a representative
spike. It has no affect on the dV/dt of the transition. switching waveform are shown in Figure 28.
ZENER SOFT SNUBBER SNUBBER SOFT ZENER
CLAMP CLAMP CLAMP CLAMP
Figure 27. Common Methods for Controlling Voltage Spikes and/or RFI
CLAMP
SNUBBER
ORIGINAL
WAVEFORM
t, TIME (sec)
Figure 28. The Effects of a Snubber versus a Clamp
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VOLTAGE (VOLTS)
SMPSRM
The Lossless Snubber 2. When the lossless snubber is reset, the
A lossless snubber is a snubber whose trapped energy energy should be returned to the input
is recovered by the power circuit. The lossless snubber is capacitor or back into the output power path.
designed to absorb a fixed amount of energy from the Study the supply carefully. Returning the
transition of a switched AC voltage node. This energy is energy to the input capacitor allows the supply
stored in a capacitor whose size dictates how much to use the energy again on the next cycle.
energy the snubber can absorb. A typical implementation Returning the energy to ground in a boost
of a lossless snubber can be seen in Figure 29. mode supply does not return the energy for
The design for a lossless snubber varies from topology reuse, but acts as a shunt current path around
to topology and for each desired transition. Some the power switch. Sometimes additional
adaptation may be necessary for each circuit. The transformer windings are used.
important factors in the design of a lossless snubber are: 3. The reset current waveform should be band
1. The snubber must have initial conditions that limited with a series inductor to prevent
allow it to operate during the desired transition additional EMI from being generated. Use of a
and at the desired voltages. Lossless snubbers 2 to 3 turn spiral PCB inductor is sufficient to
should be emptied of their energy prior to the greatly lower the di/dt of the energy exiting the
desired transition. The voltage to which it is lossless snubber.
reset dictates where the snubber will begin to
operate. So if the snubber is reset to the input
voltage, then it will act as a lossless clamp which
will remove any spikes above the input voltage.
Unsnubbed VSW
+ Snubbed VSW
VSW ID
Drain Current (ID)
Figure 29. Lossless Snubber for a One Transistor Forward or Flyback Converter
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SMPSRM
The Active Clamp stored energy) just prior to the turn off transition. It is
An active clamp is a gated MOSFET circuit that allows then disabled during the negative transition.
the controller IC to activate a clamp or a snubber circuit Obviously, the implementation of an active clamp is
at a particular moment in a switching power supply s more expensive than other approaches, and is usually
cycle of operation. An active clamp for a flyback reserved for very compact power supplies where heat is
converter is shown in Figure 30. a critical issue.
In Figure 30, the active clamp is reset (or emptied of its
Unclamped
Switch Voltage
(VSW)
Clamped Switch
Voltage (VSW)
Vin
Switch
Current (ISW)
+
ICL
VDR
+
ISW VSW
Drive
Voltage (VDR)
GND
Discharge Charge
Clamp
Current (ICL)
Figure 30. An Active Clamp Used in a One Transistor Forward or a Flyback Converter
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SMPSRM
Quasi Resonant Topologies switching frequencies are in the 100 s of kHz.
Schematically, quasi resonant topologies are minor
A quasi resonant topology is designed to reduce or
modifications of the standard PWM topologies. A
eliminate the frequency dependent switching losses
resonant tank circuit is added to the power switch section
within the power switches and rectifiers. Switching
to make either the current or the voltage ring through
losses account for about 40% of the total loss within a
a half a sinusoid waveform. Since the sinusoid starts at
PWM power supply and are proportional to the switching
zero and ends at zero, the product of the voltage and
frequency. Eliminating these losses allows the designer
current at the starting and ending points is zero, thus has
to increase the operating frequency of the switching
no switching loss.
power supply and so use smaller inductors and
There are two quasi resonant methods: zero current
capacitors, reducing size and weight. In addition, RFI
switching (ZCS) or zero voltage switching (ZVS). ZCS
levels are reduced due to the controlled rate of change of
is a fixed on time, variable off time method of control.
current or voltage.
ZCS starts from an initial condition where the power
The downside to quasi resonant designs is that they
switch is off and no current is flowing through the
are more complex than non resonant topologies due to
resonant inductor. The ZCS quasi resonant buck
parasitic RF effects that must be considered when converter is shown in Figure 31.
ILR
LR LO
CR D
VSW
Vin Vout
Cout
Cin CONTROL
FEEDBACK
A ZCS Quasi Resonant Buck Converter
SWITCH
TURN-OFF
Vin
POWER SWITCH
ON
IPK
Figure 31. Schematic and Waveforms for a ZCS Quasi-Resonant Buck Converter
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SW
V
LR
I
D
V
SMPSRM
In this design, both the power switch and the catch power delivered to the load, the amount of resonant off
diode operate in a zero current switching mode. Power is times are varied. For light loads, the frequency is high.
passed to the output during the resonant periods. So to When the load is heavy, the frequency drops. In a typical
increase the power delivered to the load, the frequency ZVS power supply, the frequency typically varies 4:1
would increase, and vice versa for decreasing loads. In over the entire operating range of the supply.
typical designs the frequency can change 10:1 over the There are other variations on the resonant theme that
ZCS supply s operating range. promote zero switching losses, such as full resonant
The ZVS is a fixed off time, variable on time method PWM, full and half bridge topologies for higher power
control. Here the initial condition occurs when the power and resonant transition topologies. For a more detailed
switch is on, and the familiar current ramp is flowing treatment, see Chapter 4 in the Power Supply
through the filter inductor. The ZVS quasi resonant buck Cookbook (Bibliography reference 2).
converter is shown in Figure 32. Here, to control the
LR LO
CR D
Vin
VI/P
Cout Vout
FEEDBACK
Cin
CONTROL
A ZVS Quasi Resonant Buck Converter
Vin
POWER SWITCH
TURNS ON
0
V * V
out
in
IPK
L ) L
R O
V
in
L
R ILOAD
0
Figure 32. Schematic and Waveforms for a
ZVS Quasi-Resonant Buck Converter
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I/P
V
SW
I
D
I
SMPSRM
requiring all electrical equipment connected to
Power Factor Correction
a low voltage distribution system to minimize
Power Factor (PF) is defined as the ratio of real power
current harmonics and maximize power factor.
to apparent power. In a typical AC power supply
2. The reflected power not wasted in the
application where both the voltage and current are
resistance of the power cord may generate
sinusoidal, the PF is given by the cosine of the phase
unnecessary heat in the source (the local
angle between the input current and the input voltage and
step down transformer), contributing to
is a measure of how much of the current contributes to
premature failure and constituting a fire hazard.
real power in the load. A power factor of unity indicates
3. Since the ac mains are limited to a finite current
that 100% of the current is contributing to power in the
by their circuit breakers, it is desirable to get
load while a power factor of zero indicates that none of
the most power possible from the given current
the current contributes to power in the load. Purely
available. This can only happen when the
resistive loads have a power factor of unity; the current
power factor is close to or equal to unity.
through them is directly proportional to the applied
The typical AC input rectification circuit is a diode
voltage.
bridge followed by a large input filter capacitor. During
The current in an ac line can be thought of as consisting
the time that the bridge diodes conduct, the AC line is
of two components: real and imaginary. The real part
driving an electrolytic capacitor, a nearly reactive load.
results in power absorbed by the load while the imaginary
This circuit will only draw current from the input lines
part is power being reflected back into the source, such
when the input s voltage exceeds the voltage of the filter
as is the case when current and voltage are of opposite
capacitor. This leads to very high currents near the peaks
polarity and their product, power, is negative.
of the input AC voltage waveform as seen in Figure 33.
It is important to have a power factor as close as
Since the conduction periods of the rectifiers are small,
possible to unity so that none of the delivered power is
the peak value of the current can be 3 5 times the average
reflected back to the source. Reflected power is
input current needed by the equipment. A circuit breaker
undesirable for three reasons:
only senses average current, so it will not trip when the
1. The transmission lines or power cord will
peak current becomes unsafe, as found in many office
generate heat according to the total current
areas. This can present a fire hazard. In three phase
being carried, the real part plus the reflected
distribution systems, these current peaks sum onto the
part. This causes problems for the electric neutral line, not meant to carry this kind of current, which
utilities and has prompted various regulations again presents a fire hazard.
Power
not used
Power used
I
110/220
+
AC VOLTS IN
DC To Power
Clarge Supply
IAV
Figure 33. The Waveforms of a Capacitive Input Filter
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VOLTAGE
CURRENT
SMPSRM
A Power Factor Correction (PFC) circuit is a switching pulses generate more heat than a purely resistive load of
power converter, essentially a boost converter with a very the same power. The active power factor correction
wide input range, that precisely controls its input current circuit is placed just following the AC rectifier bridge. An
on an instantaneous basis to match the waveshape and example can be seen in Figure 34.
phase of the input voltage. This represents a zero degrees Depending upon how much power is drawn by the unit,
or 100 percent power factor and mimics a purely resistive there is a choice of three different common control
load. The amplitude of the input current waveform is modes. All of the schematics for the power sections are
varied over longer time frames to maintain a constant the same, but the value of the PFC inductor and the
voltage at the converter s output filter capacitor. This control method are different. For input currents of less
mimics a resistor which slowly changes value to absorb than 150 watts, a discontinuous mode control scheme is
the correct amount of power to meet the demand of the typically used, in which the PFC core is completely
load. Short term energy excesses and deficits caused by emptied prior to the next power switch conduction cycle.
sudden changes in the load are supplemented by a bulk For powers between 150 and 250 watts, the critical
energy storage capacitor , the boost converter s output conduction mode is recommended. This is a method of
filter device. The PFC input filter capacitor is reduced to control where the control IC senses just when the PFC
a few microfarads, thus placing a half wave haversine core is emptied of its energy and the next power switch
waveshape into the PFC converter. conduction cycle is immediately begun; this eliminates
The PFC boost converter can operate down to about any dead time exhibited in the discontinuous mode of
30 V before there is insufficient voltage to draw any more control. For an input power greater than 250 watts, the
significant power from its input. The converter then can continuous mode of control is recommended. Here the
begin again when the input haversine reaches 30 V on the peak currents can be lowered by the use of a larger
next half wave haversine. This greatly increases the inductor, but a troublesome reverse recovery
conduction angle of the input rectifiers. The drop out characteristic of the output rectifier is encountered,
region of the PFC converter is then filtered (smoothed) which can add an additional 20 40 percent in losses to the
by the input EMI filter. PFC circuit.
A PFC circuit not only ensures that no power is Many countries cooperate in the coordination of their
reflected back to the source, it also eliminates the power factor requirements. The most appropriate
high current pulses associated with conventional document is IEC61000 3 2, which encompasses the
rectifier filter input circuits. Because heat lost in the performance of generalized electronic products. There
transmission line and adjacent circuits is proportional to are more detailed specifications for particular products
the square of the current in the line, short strong current made for special markets.
Switch Current
Input Voltage
I
Vout
Vsense
Csmall
+
Control
Clarge To Power
Supply
Conduction Angle
Figure 34. Power Factor Correction Circuit
IAVG
Figure 35. Waveform of Corrected Input
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Current
Voltage
SMPSRM
Bibliography
1. Ben Yaakov Sam, Gregory Ivensky, Passive Lossless Snubbers for High Frequency PWM Converters,
Seminar 12, APEC 99.
2. Brown, Marty, Power Supply Cookbook, Butterworth Heinemann, 1994, 2001.
3. Brown, Marty, Laying Out PC Boards for Embedded Switching Supplies, Electronic Design, Dec. 1999.
4. Martin, Robert F., Harmonic Currents, Compliance Engineering 1999 Annual Resources Guide, Cannon
Communications, LLC, pp. 103 107.
5. ON Semiconductor, Rectifier Applications Handbook, HB214/D, Rev. 2, Nov. 2001.
http://onsemi.com
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SMPSRM
SWITCHMODE Power Supply Examples
This section provides both initial and detailed information to simplify the selection and design of a variety of
SWITCHMODE power supplies. The ICs for Switching Power Supplies figure identifies control, reference voltage,
output protection and switching regulator ICs for various topologies.
Page
ICs for Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Integrated circuits identified for various sections of a switching power supply.
Suggested Components for Specific Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
A list of suggested control ICs, power transistors and rectifiers for SWITCHMODE power supplies by application.
" CRT Display System . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
" AC/DC Power Supply for CRT Displays . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
" AC/DC Power Supply for Storage, Imaging & Entertainment . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
" DC DC Conversion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
" Typical PC Forward Mode SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
Real SMPS Applications
80 W Power Factor Correction Controller . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
Compact Power Factor Correction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
Monitor Pulsed Mode SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
70 W Wide Mains TV SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
100 W Wide Mains TV SMPS with 1.3 W Stand by . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
Low Cost Off line IGBT Battery Charger . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50
110 W Output Flyback SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
Efficient Safety Circuit for Electronic Ballast . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
AC DC Battery Charger Constant Current with Voltage Limit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 55
Some of these circuits may have a more complete application note, spice model information or even an evaluation board
available. Consult ON Semiconductor s website (http://onsemi.com) or local sales office for more information.
http://onsemi.com
35
SNUBBER/
CLAMP
1N62xxA OUTPUT FILTERS
POWER MOS
1N63xxA
DRIVERS DC DC CONVERSION
MBR1100 MBRS240L
MUR160
MBR3100 MBRS360
MUR260
MC33151
CS51031
MC34063A
MURS160 MBR360 MURHF860CT
CS51033
MC34163
MC33152
MURS260
MBRD360 MURS360
CS51411
MC34166
MC33153
P6KExxxA
POWER FACTOR MBRS1100 CS51412
MC34167
P6SMB1xxA
CORRECTION
CS51413
NCP1400A
OUTPUT PROTECTION
CS51414
NCP1402
MC33260
CS5171
NCP1410
MC33262 MAX707 MC33161
CS5172
NCP1411
POWER MOS
MC33368 MAX708 MC33164
CS5173 NCP1417
DRIVERS
MC34262 MAX809 MC3423
NCP1450A
CS5174
NCP1650
MAX810 NCP30x MC33463 NCP1550
NCP1651 MC33466
MC33064 NCP803
POWER FACTOR SNUBBER/ OUTPUT OUTPUT VOLTAGE
TRANS
CORRECTION CLAMP FILTERS PROTECTION REGULATION
FORMERS
DC DC
CONVERSION
Vref
POWER
SWITCH
PWM
L4949
MC33275 MC7905A
REF
OSC
LM2931
MC33761 MC7906
LM2935
MC34268 MC7908
STARTUP CONTROL
LM317
MC78xx MC7908A
VOLTAGE
FEEDBACK LM317L
MC78Bxx MC7912
LM317M
MC78Fxx MC7915
LM337
MC78Lxx MC7918
MC33362
CS5101
LM350
MC78Mxx MC7924
MC33363
MC44604
CS3843 CS51227 NCP100
LP2950
MC33365 MC78PCxx MC79Mxx
MC44605
CS51021 CS5124
TL431/A/B
NCP100x LP2951
MC78Txx NCP1117
MC33023 MC44608
CS51022
TLV431A
NCP105x MC33263 MC7905 NCP50x
MC33025 NCP1200
CS51023
MMBZ52xx
MC33065 NCP1205 MC33269 MC7905.2 NCP51x
CS51024
VOLTAGE
HV SWITCHING
MMSZ52xx
UC384x
MC33067
CS5106
FEEDBACK
REGULATORS
MMSZ46xx
MC33364 VOLTAGE REGULATION
CS51220
MC44603A
CS51221
STARTUP
CONTROL
Figure 30. Integrated Circuits for Switching Power Supplies
SMPSRM
Figure 36. . Intergrated Ciruits
for Switching Power Supplies
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Video
RWM
On Screen Display
Monitor Driver
I2C BUS
Generator
MCU 10101100101
R
Overlayed
RGB
RGB
V_Sync
SYNC PROCESSOR
G
CRT
V_Sync
H_Sync
1280
x
1024
H_Sync
B
RGB
HC05
CPU
MEMORY
CORE
PWM
R
or I2C
Vertical
G Driver
RGB
B
DOWN
USB HUB
H Driver TR
MTD6N10/15
H Driver
Geometry Correction
UP
MC33363A/B
Line Driver
NCP100x
NCP105x
USB & Auxiliary Standby
NCP1200
IRF630 / 640 / 730 /740 / 830 / 840
AC/DC
Damper Diode
Power Supply
Timebase Processor
MUR8100E
V_Sync
DC TO DC
H Output TR
MUR4100E
CONTROLLER
Line
H_Sync
MUR460
A.C.
600V 8A
UC3842/3
PFC Devices
N Ch
MTP6P20E
NCP1650
S.M.P.S MOSFET
NCP1651
Controller
MC34262
MC33368
MC33260
UC384x
MUR420
Sync
MC44603/5
MUR440
Signal
MC44608
MUR460
NCP1200
NCP1205
Figure 31. 15 Monitor Power Supplies
Figure 37.
Supplies
Monitor
Power
. 15
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SMPSRM
SMPSRM
Ultrafast
Start up
+
Rectifier
Switch
Load
Rectifier
+
Bulk
Storage
Capacitor
PWM
MOSFET
Control
IC
AC
n outputs
Line
Prog.
Prec.
Ref
PWM Switcher
Figure 38. AC/DC Power Supply for CRT Displays
Table 1.
Part # Description Key Parameters Samples/Prod.
MC33262 PFC Control IC Critical Conduction PFC Controller Now/Now
MC33368 PFC Control IC Critical Conduction PFC Controller + Internal Start up Now/Now
MC33260 PFC Control IC Low System Cost, PFC with Synchronization Now/Now
Capability, Follower Boost Mode, or Normal Mode
MC33365 PWM Control IC Fixed Frequency Controller + 700 V Start up, 1 A Now/Now
Power Switch
MC33364 PWM Control IC Variable Frequency Controller + 700 V Start up Switch Now/Now
MC44603A/604 PWM Control IC GreenLine, Sync. Facility with Low Standby Mode Now/Now
MC44605 PWM Control IC GreenLine, Sync. Facility, Current mode Now/Now
MC44608 PWM Control IC GreenLine, Fixed Frequency (40 kHz, 75 kHz and 100 Now/Now
kHz options), Controller + Internal Start up, 8 pin
MSR860 Ultrasoft Rectifier 600 V, 8 A, trr = 55 ns, Ir max = 1 uA Now/Now
MUR440 Ultrafast Rectifier 400 V, 4 A, trr = 50 ns, Ir max = 10 uA Now/Now
MRA4006T3 Fast Recovery Rectifier 800 V, 1 A, Vf = 1.1 V @ 1.0 A Now/Now
MR856 Fast Recovery Rectifier 600 V, 3 A, Vf = 1.25 V @ 3.0 A Now/Now
NCP1200 PWM Current Mode Controller 110 mA Source/Sink, O/P Protection, 40/60/110 kHz Now/Now
NCP1205 Single Ended PWM Controller Quasi resonant Operation, 250 mA Source/Sink, Now/Now
8 36 V Operation
UC3842/3/4/5 High Performance Current Mode 500 kHz Freq., Totem Pole O/P, Cycle by Cycle Now/Now
Controllers Current Limiting, UV Lockout
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SMPSRM
Ultrafast
Start up
+
Rectifier
Switch
Load
Rectifier
+
Bulk
Storage
Capacitor
PWM
MOSFET
Control
IC
AC
n outputs
Line
Prog.
Prec.
Ref
PWM Switcher
Figure 39. AC/DC Power Supply for Storage,
Imaging & Entertainment
Table 2.
Part # Description Key Parameters Samples/Prod.
MC33363A/B/65 PWM Control IC Controller + 700 V Start up & Power Switch, < 15 W Now/Now
MC33364 PWM Control IC Critical Conduction Mode, SMPS Controller Now/Now
TL431B Program Precision Reference 0.4% Tolerance, Prog. Output up to 36 V, Temperature Now/Now
Compensated
MSRD620CT Ultrasoft Rectifier 200 V, 6 A, trr = 55 ns, Ir max = 1 uA Now/Now
MR856 Fast Recovery Rectifier 600 V, 3 A, Vf = 1.25 V @ 3.0 A Now/Now
NCP1200 PWM Current Mode Controller 110 mA Source/Sink, O/P Protection, 40/60/110 kHz Now/Now
NCP1205 Single Ended PWM Controller Quasi resonant Operation, 250 mA Source/Sink, Now/Now
8 36 V Operation
UC3842/3/4/5 High Performance Current Mode 500 kHz Freq., Totem Pole O/P, Cycle by Cycle Now/Now
Controllers Current Limiting, UV Lockout
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39
SMPSRM
Lo Lo
Voltage
Regulation
+ +
+ +
Vin Co Vout Load Vin Co Vout Load
Control IC
Control IC
Buck Regulator Synchronous Buck Regulator
Figure 40. DC DC Conversion
Table 3.
Part # Description Key Parameters Samples/Prod.
MC33263 Low Noise, Low Dropout 150 mA; 8 Outputs 2.8 V 5 V; SOT 23L 6 Lead Now/Now
Regulator IC Package
MC33269 Medium Dropout Regulator IC 0.8 A; 3.3; 5, 12 V out; 1 V diff; 1% Tolerance Now/Now
MC33275/375 Low Dropout Regulator 300 mA; 2.5, 3, 3.3, 5 V out Now/Now
LP2950/51 Low Dropout, Fixed Voltage IC 0.1 A; 3, 3.3, 5 V out; 0.38 V diff; 0.5% Tolerance Now/Now
MC78PC CMOS LDO Linear Voltage Iout = 150 mA, Available in 2.8 V, 3 V, 3.3 V, 5 V; SOT Now/Now
Regulator 23 5 Leads
MC33470 Synchronous Buck Regulator IC Digital Controlled; Vcc = 7 V; Fast Response Now/Now
NTMSD2P102LR2 P Ch FET w/Schottky in SO 8 20 V, 2 A, 160 mW FET/1 A, Vf = 0.46 V Schottky Now/Now
NTMSD3P102R2 P Ch FET w/Schottky in SO 8 20 V, 3 A, 160 mW FET/1 A, Vf = 0.46 V Schottky Now/Now
MMDFS6N303R2 N Ch FET w/Schottky in SO 8 30 V, 6 A, 35 mW FET/3 A, Vf = 0.42 V Schottky Now/Now
NTMSD3P303R2 P Ch FET w/Schottky in SO 8 30 V, 3 A, 100 mW FET/3 A, Vf = 0.42 V Schottky Now/Now
MBRM140T3 1A Schottky in POWERMITEŁł 40 V, 1 A, Vf = 0.43 @ 1 A; Ir = 0.4 mA @ 40 V Now/Now
Package
MBRA130LT3 1A Schottky in SMA Package 40 V, 1 A, Vf = 0.395 @ 1 A; Ir = 1 mA @ 40 V Now/Now
MBRS2040LT3 2A Schottky in SMB Package 40 V, 2 A, Vf = 0.43 @ 2 A; Ir = 0.8 mA @ 40 V Now/Now
MMSF3300 Single N Ch MOSFET in SO 8 30 V, 11.5 A(1), 12.5 mW @ 10 V Now/Now
NTD4302 Single N Ch MOSFET in DPAK 30 V, 18.3 A(1), 10 mW @ 10 V Now/Now
NTTS2P03R2 Single P Ch MOSFET in 30 V, 2.7 A, 90 mW @ 10 V Now/Now
Micro8ęł Package
MGSF3454X/V Single N Ch MOSFET in 30 V, 4.2 A, 65 mW @ 10 V Now/Now
TSOP 6
NTGS3441T1 Single P Ch MOSFET in 20 V, 3.3 A, 100 mW @ 4.5 V Now/Now
TSOP 6
NCP1500 Dual Mode PWM Linear Buck Prog. O/P Voltage 1.0, 1.3, 1.5, 1.8 V Now/Now
Converter
NCP1570 Low Voltage Synchronous Buck UV Lockout, 200 kHz Osc. Freq., 200 ns Response Now/Now
Converter
NCP1571 Low Voltage Synchronous Buck UV Lockout, 200 kHz Osc. Freq., 200 ns Response Now/Now
Converter
CS5422 Dual Synchronous Buck 150 kHz 600 kHz Prog. Freq., UV Lockout, 150 ns Now/Now
Converter Transient Response
(1) Continuous at TA = 25 C, Mounted on 1 square FR 4 or G10, VGS = 10 V t 10 seconds
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40
Part No. VRRM (V) I (A) Package
Part No. VRRM (V) I (A) Package
o
o
MBR2535CTL 35 25 TO 220
MBR160 60 1 Axial
Part No. VRRM (V) I (A) Package
o
+3.3 V 14 A
1N5404RL 400 1000 3 Axial
X
1N5406RL 400 1000 3 Axial
X
+
Part No. VRRM (V) I (A) Package
o
1N5408RL 400 1000 3 Axial
X
MBR2535CTL 35 25 TO 220
MBR2545CT 45 25 TO 220
MBR3045ST 45 30 TO 220
+5 V 22 A
MBRF2545CT 45 25 TO 220
+
Mains
+
MBR3045PT 45 30 TO 218
230 Vac
Voltage
MBR3045WT 45 30 TO 247
Stand by
5 V 0.1 A
Part No. VRRM (V) I (A) Package
o
+12 V 6 A
MBR2060CT 60 20 TO 220
MBR20100CT 100 20 TO 220
+
MBR20200CT 200 20 TO 220
MUR1620CT 200 16 TO 220
MUR1620CTR 200 16 TO 220
5 V 0.5 A
MURF1620CT 200 16 TO 220
Part No. Package
I (A) Package
Part No. VRRM (V)
U384X Series DIP8/SO 8/SO 14 o
+
MC34060 DIP14/SO 14
MBRS340T3 40 3 SMC
PWM
TL494 DIP16/SO 16
MBRD340 40 3 DPAK
IC
TL594 DIP16/SO 16
1N5821 30 3 Axial
12 V 0.8 A
MC34023 DIP16/SO 16
1N5822 40 3 Axial
MC44608 DIP8
MBR340 40 3 Axial
+
MC44603 DIP16/SO 16
MC44603A DIP16/SO 16
Part No. VRRM (V) I (A) Package
o
MBR3100 100 3 Axial
MATRIX
Part No. Package
TL431 TO 92
Part No. VRRM (V) I (A) Package
o
MUR180E, MUR1100E 600 1000 1 Axial
X
X
MUR480E, MUR4100E 600 1000 4 Axial
X
MR756RL, MR760RL 600 1000 6 Axial
1N4937 600 1 Axial
Figure 35. Typical 200 W ATX Forward Mode SMPS
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41
ATX Forward Mode SMPS
Figure 41. . Typical 200 W
SMPSRM
SMPSRM
Application: 80 W Power Factor Controller
100 k
1
C5
R6 1N4934
MC33262
D6
8
D2 D4
+
100
+ 36 V
ZERO CURRENT
1.2 V
C4
T
DETECTOR
92 to
D1 D3 6.7 V 5 22 k
+
138 Vac
1.6 V/
R4
1.4 V
UVLO
2.5 V
13 V/
REFERENCE +
MUR130
8.0 V
D5 VO
16 V
TIMER R 230 V/
500 V/8 A
10
0.35 A
7
N Ch
DRIVE
DELAY +
MOSFET
OUTPUT
220
10
RS Q1
C3
LATCH
2.2 M 1.0 M
20 k 4
R5 R2
1.5 V 0.1
OVERVOLTAGE
10 pF
R7
COMPARATOR
CURRENT
SENSE
+
1.08 Vref
COMPARATOR
ERROR AMP
+
10 mA
Vref 1
MULTIPLIER
0.01 7.5 k
3
11 k
C2 R3
QUICKSTART
R1
2
6
0.68
C1
Figure 42. 80 W Power Factor Controller
Features:
Reduced part count, low cost solution.
ON Semiconductor Advantages:
Complete semiconductor solution based around highly integrated MC33262.
Devices:
Part Number Description
MC33262 Power Factor Controller
MUR130 Axial Lead Ultrafast Recovery Rectifier (300 V)
Transformer Coilcraft N2881 A
Primary: 62 turns of #22 AWG
Secondary: 5 turns of #22 AWG
Core: Coilcraft PT2510
Gap: 0.072 total for a primary inductance (Lp) of 320 mH
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42
RFI
FILTER
SMPSRM
Application: Compact Power Factor Correction
Vcc
FUSE
0.33 F
1N5404
L1
+
10 F/
MAINS
100 nF
16 V
AC LINE Vout
MUR460
FILTER
+
100 F/
450 V
1 8
500 V/8 A
N Ch
2 7
100 nF
MOSFET
10 W
3 6
4 5
12 kW
1 MW
120 pF
45 kW 1 MW
0.5 W/3 W
Figure 43. Compact Power Factor Correction
Features :
Low cost system solution for boost mode follower.
Meets IEC1000 3 2 standard.
Critical conduction, voltage mode.
Follower boost mode for system cost reduction smaller inductor and MOSFET can be used.
Inrush current detection.
Protection against overcurrent, overvoltage and undervoltage.
ON Semiconductor advantages:
Very low component count.
No Auxiliary winding required.
High reliability.
Complete semiconductor solution.
Significant system cost reduction.
Devices:
Part Number Description
MC33260 Power Factor Controller
MUR460 Ultrafast Recovery Rectifier (600 V)
1N5404 General Purpose Rectifier (400 V)
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43
MC33260
SMPSRM
Application: Monitor Pulsed Mode SMPS
90 Vac to
270 Vac
22 H
1 nF/1 kV
RFI
90 V/0.1 A
FILTER
MR856
+ +
1 nF/500 V
4.7 MW
47 F 47 F
1 W
Vin 120 pF
D1 D4
1N5404
150 F
3.9 kW/6 W
4.7 kW
400 V
1N4148
1N4934 MCR22 6
1 nF/500 V
2W
100 nF
1N4934
22 kW
SYNC 45 V/
MR856
+ +
47 F
1 A
25 V
MR856 1000 F
3.3 kW
1 H
1.2 kW 47 kW
10 pF
SMT31
9 8
2.2 nF
10 7
15 V/
MR852
+
470 pF
0.8 A
4.7 F
2.2 kW
11 6
1000 F
+
4.7 F+
Lp
8.2 kW
10 V
12 5
22
470 4.7 F+ 150 kW
1N4148 560 kW
nF
kW 10 V
13 4
470 pF
10 V/
MR852
+
2.2 nF
0.3 A
Note 1
14 3
56 kW
220 F
10 W
1N4934
15 2
1 kW
270 W
16 1
8 V/
MBR360
470 W
+
1.5 A
0.1 W
56 kW
10 kW
4700 F
100 W
MOC8107
1.8 MW
96.8 kW
10 kW
Vin
100 nF
TL431
1N4742A
2.7 kW
12 V
2.7 kW
Note 1: 500 V/8 A N Channel MOSFET
1 kW
100 nF
BC237B
VmP
FROM mP
0: STAND BY
1: NORMAL MODE
Figure 44. Monitor Pulsed Mode SMPS
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44
MC44605P
SMPSRM
Features:
Off power consumption: 40 mA drawn from the 8 V output in Burst mode.
Vac (110 V) ł about 1 watt
Vac (240 V) ł about 3 watts
Efficiency (pout = 85 watts)
Around 77% @ Vac (110 V)
Around 80% @ Vac (240 V)
Maximum Power limitation.
Over temperature detection.
Winding short circuit detection.
ON Semiconductor Advantages:
Designed around high performance current mode controller.
Built in latched disabling mode.
Complete semiconductor solution.
Devices:
Part Number Description
MC44605P High Safety Latched Mode GreenLinet Controller
For (Multi) Synchronized Applications
TL431 Programmable Precision Reference
MR856 Fast Recovery Rectifier (600 V)
MR852 Fast Recovery Rectifier (200 V)
MBR360 Axial Lead Schottky Rectifier (60 V)
BC237B NPN Bipolar Transistor
1N5404 General Purpose Rectifier (400 V)
1N4742A Zener Regulator (12 V, 1 W)
Transformer G6351 00 (SMT31M) from Thomson Orega
Primary inductance = 207 mH
Area = 190 nH/turns2
Primary turns = 33
Turns (90 V) = 31
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45
SMPSRM
Application: 70 W Wide Mains TV SMPS
95 Vac to
265 Vac
F1
FUSE 1.6 A
C30
100 nF
250 Vac
RFI
LF1
FILTER
C19
1 nF/1 kV
R21
4.7 MW
D1 D4
1N4007
C1
R16
220 mF
L3
3.8 MW 68 kW/2 W C15 220 pF
22 H
115 V/0.45 A
C4 C5
1 nF/1 kV R7 D13
C26
D12
68 kW/1 W 1N4148
4.7 nF
C20 D23
MR856
47 F 47 F
C16
D15
100 F
1N4148
D7
L1
1N4937
1 H
R19
C9
15 V/1.5 A
27 kW
100 nF
9 8
D5
C11
C21
C8 560 pF
R22
MR854
100 pF
10 7 1000 F
C12
1 kW
C10 1 F
R3
1 nF
11 6
22 kW
R18 15 kW
11 V/0.5 A
12 5
5.6 kW
180 kW
D8
Q1
C22
13 4
MR854
C7 600 V/4 A
1000 F
R15
10 nF N Ch
1 MW
14 3
MOSFET
R8
R20 47W
OREGA TRANSFORMER
1 kW
15 2
G6191 00
R4
R9 150W
THOMSON TV COMPONENTS
3.9 kW
16 1
C14
R5
R33
220 pF
2.2 kW
0.31 W
R14 R13
47 kW 10 kW
Figure 45. 70 W Wide Mains TV SMPS
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46
MC44603AP
SMPSRM
Features:
70 W output power from 95 to 265 Vac.
Efficiency
@ 230 Vac = 86%
@ 110 Vac = 84%
Load regulation (115 Vac) = " 0.8 V.
Cross regulation (115 Vac) = " 0.2 V.
Frequency 20 kHz fully stable.
ON Semiconductor Advantages:
DIP16 or SO16 packaging options for controller.
Meets IEC emi radiation standards.
A narrow supply voltage design (80 W) is also available.
Devices:
Part Number Description
MC44603AP Enhanced Mixed Frequency Mode
GreenLinet PWM Controller
MR856 Fast Recovery Rectifier (600 V)
MR854 Fast Recovery Rectifier (400 V)
1N4007 General Purpose Rectifier (1000 V)
1N4937 General Purpose Rectifier (600 V)
Transformer Thomson Orega SMT18
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47
SMPSRM
Application: Wide Mains 100 W TV SMPS with 1.3 W TV Stand by
F1
C31
100 nF
C19
RFI
47283900 R F6
2N2F Y
FILTER
C3
R16 4.7 MW/4 kV
1 nF
C11
D1 D4
220 pF/500 V
1N5404
+ C5
112 V/0.45 A
R1
220 mF 1
14
22 kW
400 V
+
C4 D18 MR856
5W
J3
1 nF
2
C12 C13
C6
12 47 F/250 V 100 nF
D5 47 nF
16 V/1.5 A
3
1N4007 630 V
D6
R7 47 k&! C17 120 pF
R5 100 kW
MR856
1
6 1
J4
D12 DZ1
D7
2
1N4934 MCR22 6
1N4148
11
8 V/1 A
7
1 8 2
D9 MR852
3
+
C14
+ C7
Isense
1000 F/35 V
22 mF
2 7
10
16 V
Vcc
C16 R19
3 6 C9
D13
100 pF 18 kW
470 pF
1N4148
630 V
4 5
R2 600 V/6 A
10 W N CH 8
D14
D10
MOSFET
+
R12 C18
C8
MR856
MR852
C15
1 kW 100 nF
100 nF
1000 F/16 V
9
R17
2.2 kW
5 W
R4 3.9 kW
R3
0.27 W
ON OFF
R21 47 W
R9
100 kW
OPT1
ON = Normal mode
R10
OFF = Pulsed mode
10 kW
C19
33 nF
R11 DZ3
47 kW 10 V
DZ2
R8
1N4740A
TL431CLP
2.4 kW
Figure 46. Wide Mains 100 W TV SMPS with Secondary
Reconfiguration for 1.3 W TV Stand by
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48
MC44608P75
SMPSRM
Features:
Off power consumption: 300mW drawn from the 8V output in pulsed mode.
Pin = 1.3W independent of the mains.
Efficiency: 83%
Maximum power limitation.
Over temperature detection.
Demagnetization detection.
Protection against open loop.
ON Semiconductor Advantages:
Very low component count controller.
Fail safe open feedback loop.
Programmable pulsed mode power transfer for efficient system stand by mode.
Stand by losses independent of the mains value.
Complete semiconductor solution.
Devices:
Part Number Description
MC44608P75 GreenLinet Very High Voltage PWM Controller
TL431 Programmable Precision Reference
MR856 Fast Recovery Rectifier (600 V)
MR852 Fast Recovery Rectifier (200 V)
1N5404 General Purpose Rectifier (400 V)
1N4740A Zener Regulator (10 V, 1 W)
Transformer SMT19 40346 29 (9 slots coil former)
Primary inductance: 181 mH
Nprimary: 40 turns
N 112 V: 40 turns
N 16 V: 6 turns
N 8 V: 3 turns
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SMPSRM
Application: Low Cost Offline IGBT Battery Charger
+
130 to 350 V DC
C3
R1 D1
8 V at 400 mA
+ +
220 mF/
C2
150 10 V
1N4148
220 mF/
D3
10 V
R3
C10
R13
MBRS240LT3
220 k
D4
1 nF
100 k
D5
R2
1N4937
150
1N4148
M1
R1
R11
+
MMG05N60D
113 k
120 k
C3
R5
10 mF/
IC1
350 V
1 k MOC8103
MC14093
R5
1.2 k
8 7 6 5
+
C8
MC33341
1 mF
+
1 2 3 4 D4
C7
R9
C4 12 V
10 mF
47 nF
470
R12
R2
D2 C5
20 k
R9
Q1 C9
3.9
12 V 1 nF
Q5
100
MBT3946DW 1 nF
R10
0 V
Figure 47. Low Cost Offline IGBT Battery Charger
Features:
Universal ac input.
3 Watt capability for charging portable equipment.
Light weight.
Space saving surface mount design.
ON Semiconductor Advantages:
Special process IGBT (Normal IGBTs will not function properly in this application).
Off the shelf components.
SPICE model available for MC33341.
Devices:
Part Number Description
MMG05N60D Insulated Gate Bipolar Transistor in SOT 223 Package
MC33341 Power Supply Battery Charger Regulator Control Circuit
MBT3946DW Dual General Purpose (Bipolar) Transistors
MBRS240LT3 Surface Mount Schottky Power Rectifier
MC14093 Quad 2 Input NAND Schmitt Trigger
1N4937 General Purpose Rectifier (600 V)
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SMPSRM
Application: 110 W Output Flyback SMPS
180 VAC TO 280 VAC
C3
1 nF / 1 KV
RFI
R1
FILTER
1 W / 5 W
R3
C4 C7
4.7 kW
1 nF / 1000 V
C32 220 pF
120 V / 0.5 A
C1
R20
D1 D4
100 mF
22 kW
D8
D5
1N4007
5 W
MR856
1N4934
C30 C31
C2 100 mF 0.1 mF
C17
220 mF
47 nF
R2
D7
L1
68 kW / 2 W
C29 220 pF
R4 MR856
1 mH
27 kW 28 V / 1 A
Laux
9 8
R9
D6
C16
D9
1 kW
1N4148
C9 820 pF 100 pF
MR852
10 7
C27 C28
R5
C15
1000 mF 0.1 mF
1.2 kW
1 nF
C10 1 mF
11 6
LP
R7
C14
180 kW
4.7 nF
12 5
R6
C26 220 pF
R15
R8
180 W
10 kW
15 kW
15 V / 1 A
13 4
C11
Note 1
D10
1 nF
14 3
MR852
R10
R26
C25 C24
R16
10 W
1 kW
1000 mF 0.1 mF
15 2
10 kW
16 1
C23 220 pF
R18 R19 R14
8 V / 1 A
C13
27 kW 10 kW 2 X 0.56 W//
100 nF
D11
MR852
C21 C22
1000 mF 0.1 mF
R17
10 kW R24
R23
270 W
117.5 kW
R21
C19
D14
10 kW
100 nF
1N4733
C20
R25 C12 33 nF
TL431
1 kW 6.8 nF
Note 1: 600 V/ 6 A N Channel MOSFET R22
2.5 kW
Figure 48. 110 W Output Flyback SMPS
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51
MC44603P
SMPSRM
Features:
Off line operation from 180 V to 280 Vac mains.
Fixed frquency and stand by mode.
Automatically changes operating mode based on load requirements.
Precise limiting of maximum power in fixed frequency mode.
ON Semiconductor Advantages:
Built in protection circuitry for current limitation, overvoltage detection, foldback, demagnetization and softstart.
Reduced frequency in stand by mode.
Devices:
Part Number Description
MC44603P Enhanced Mixed Frequency Mode GreenLinet PWM Controller
MR856 Fast Recovery Rectifier (600 V)
MR852 Fast Recovery Rectifier (200 V)
TL431 Programmable Precision Reference
1N4733A Zener Voltage Regulator Diode (5.1 V)
1N4007 General Purpose Rectifier (1000 V)
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SMPSRM
Application: Efficient Safety Circuit for Electronic Ballast
C13 100 nF C14 100 nF
AGND
250 V 250 V
C12 22 nF
R18 PTC
C11 4.7 nF
1200 V
PTUBE =
55 W
T1A
L1 1.6 mH
FT063
Q3
Q2
MJE18004D2
MJE18004D2
R13 R14
2.2 R 2.2 R
R11
C9 R12
C8
4.7 R
2.2 nF 4.7 R
2.2 nF
DIAC
C6 10 nF C7 10 nF
NOTES: * All resistors are ą 5%, 0.25 W
D4
unless otherwise noted
* All capacitors are Polycarbonate, 63 V,
R10
T1B
ą 10%, unless otherwise noted
T1C
10 R
D3 1N4007
C5 0.22 mF
R9
330 k
C4 47 mF
+
450 V
R7 1.8 M P1 20 k
C15 100 nF
Q1
500 V/4 A N Ch 1N5407 1N5407
D2 MUR180E R6 1.0 R
MOSFET
D8 D9
3 1
C16
630 V
2 47 nF
R5 1.0 R
1N5407
T2
AGND
D7 D6
R4 22 k
7
1N5407
5 4
FILTER
+ C3 1.0 mF
2
C2
D1
6
330 mF
C17 47 nF
MUR120
25 V
1
630 V
8
R3
3
C1 10 nF
100 k/1.0 W
R2 1.2 M
FUSE
LINE
R1 12 k
220 V
Figure 49. Efficient Safety Circuit for Electronic Ballast
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53
U1
MC34262
SMPSRM
Features:
Easy to implement circuit to avoid thermal runaway when fluorescent lamp does not strike.
ON Semiconductor Advantages:
Power devices do not have to be oversized lower cost solution.
Includes power factor correction.
Devices:
Part Number Description
MC34262 Power Factor Controller
MUR120 Ultrafast Rectifier (200 V)
MJE18004D2 High Voltage Planar Bipolar Power Transistor (100 V)
1N4007 General Purpose Diode (1000 V)
1N5240B Zener Voltage Regulator Diode (10 V)
1N5407 Rectifier (3 A, 800 V)
*Other Lamp Ballast Options:
1, 2 Lamps 3, 4 Lamps
825 V BUL642D2 BUL642D2
100 V MJD18002D2 MJB18004D2
1200 V MJD18202D2 MJB18204D2
MJE18204D2
ON Semiconductor s H2BIP process integrates a diode and bipolar transistor for a single package solution.
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SMPSRM
Application: AC DC Battery Charger Constant Current with
Voltage Limit
T0.2x
J1
D1
F1
1 250R
1N4140
+ R4 5 V
2
C1 + 4 kW
LINE D8 C5
R8
10 mF/350 V
D9
J2
100
1 mF
10 V R1 1N4140 T1
BZX84/5 V
5 6
1
100 mF +
220 D3
4
2
R2
D7
C5
+
7
R5
3
4 kW
D2 C2 20 mF
MURS320T3
D4 47 k
R14
R6
BZX84/18V 22 k
1N4140
47 k
C4
R3
U2 8 7 6 5
1 nF
D6 2
22 k
8 7 1
U1
1N4140 D5 MURS160T3
Line VCC ICD
MC33341
Q1
R4
6
MC33364 R13
600 V/1 A
5
GND 12 k
330 N Ch MOSFET
2
C3
R7 1 2 3 4
Vref FL
2.7
C7
4 3
33 nF
C3 1SO1 R12
R10
5 2
100 nF 10 k
R11
100 R
MOC0102
1 0.25
4
Figure 50. AC DC Battery Charger Constant Current with Voltage Limit
Features:
Universal ac input.
9.5 Watt capability for charging portable equipment.
Light weight.
Space saving surface mount design.
ON Semiconductor Advantages:
Off the shelf components
SPICE model available for MC33341
Devices:
Part Number Description
MC33341 Power Supply Battery Charger Regulator Control Circuit
MC33364 Critical Conduction SMPS Controller
MURS160T3 Surface Mount Ultrafast Rectifier (600 V)
MURS320T3 Surface Mount Ultrafast Rectifier (200 V)
BZX84C5V1LT1 Zener Voltage Regulator Diode (5.1 V)
BZX84/18V Zener Voltage Regulator Diode (MMSZ18T1)
Transformer For details consult AN1600
http://onsemi.com
55
CC
DO
V
CSI
CTA
CMP
CSI
GND
VSI
SMPSRM
Literature Available from ON Semiconductor
Application Notes
These older Application Notes may contain part numbers that are no longer available, but the applications information
may still be helpful in designing an SMPS. They are available through the Literature Distribution Center for
ON Semiconductor at 800 344 3860 or 303 675 2175 or by email at ONlit@hibbertco.com.
AN873 Understanding Power Transistor Dynamic Behavior: dv/dt Effects on Switching RBSOA
AN875 Power Transistor Safe Operating Area: Special Consideration for Switching Power Supplies
AN913 Designing with TMOS Power MOSFETs
AN915 Characterizing Collector to Emitter and Drain to Source Diodes for Switchmode Applications
AN918 Paralleling Power MOSFETs in Switching Applications
AN920 Theory and Applications of the MC34063 and mA78S40 Switching Regulator Control Circuits
AN929 Insuring Reliable Performance from Power MOSFETs
AN952 Ultrafast Recovery Rectifiers Extend Power Transistor SOA
AN1040 Mounting Considerations for Power Semiconductors
AN1043 SPICE Model for TMOS Power MOSFETs
AN1080 External Sync Power Supply with Universal Input Voltage Range for Monitors
AN1083 Basic Thermal Management of Power Semiconductors
AN1090 Understanding and Predicting Power MOSFET Switching Behavior
AN1320 300 Watt, 100 kHz Converter Utilizes Economical Bipolar Planar Power Transistors
The following Application Notes are available directly from the ON Semiconductor website
(http://onsemi.com).
AN1327 Very Wide Input Voltage Range, Off line Flyback Switching Power Supply
AN1520 HDTMOS Power MOSFETs Excel in Synchronous Rectifier Applications
AN1541 Introduction to Insulated Gate Bipolar Transistor
AN1542 Active Inrush Current Limiting Using MOSFETs
AN1543 Electronic Lamp Ballast Design
AN1547 A DC to DC Converter for Notebook Computers Using HDTMOS and Synchronous Rectification
AN1570 Basic Semiconductor Thermal Measurement
AN1576 Reduce Compact Fluorescent Cost with Motorola s (ON Semiconductor) IGBTs for Lighting
AN1577 Motorola s (ON Semiconductor) D2 Series Transistors for Fluorescent Converters
AN1593 Low Cost 1.0 A Current Source for Battery Chargers
AN1594 Critical Conduction Mode, Flyback Switching Power Supply Using the MC33364
AN1600 AC DC Battery Charger Constant Current with Voltage Limit
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56
SMPSRM
Literature Available from ON Semiconductor (continued)
AN1601 Efficient Safety Circuit for Electronic Ballast
AN1628 Understanding Power Transistors Breakdown Parameters
AN1631 Using PSPICE to Analyze Performance of Power MOSFETs in Step Down, Switching Regulators
Employing Synchronous Rectification
AN1669 MC44603 in a 110 W Output SMPS Application
AN1679 How to Deal with Leakage Elements in Flyback Converters
AN1680 Design Considerations for Clamping Networks for Very High Voltage Monolithic Off Line PWM
Controllers
AN1681 How to Keep a Flyback Switch Mode Supply stable with a Critical Mode Controller
Brochures and Data Books
Thermal Modeling & Management of Discrete Surface Mount Packages BR1487/D
Analog/Interface ICs Device DL128/D
Bipolar Device Data DL111/D
Thyristor Device Data DL137/D
Power MOSFETs DL135/D
TVS/Zener Device Data DL150/D
Rectifier Device Data DL151/D
Master Components Selector Guide SG388/D
Device Models
Device models for SMPS circuits (MC33363 and MC33365), power transistors, rectifiers and other discrete products
are available through ON Semiconductor s website or by contacting your local sales office.
http://onsemi.com
57
SMPSRM
Reference Books Relating to Switching Power Supply Design
Baliga, B. Jayant,
Power Semiconductor Devices, PWS Publishing Co., Boston, 1996. 624 pages.
Brown, Marty,
Practical Switching Power Supply Design, Academic Press, Harcourt Brace Jovanovich, 1990. 240 pages.
Brown, Marty
Power Supply Cookbook, EDN Series for Design Engineers, ON Semiconductor Series in Solid State Electronics,
Butterworth Heinmann, MA, 1994. 238 pages
Chrysiss, G. C.,
High Frequency Switching Power Supplies: Theory and Design, Second Edition, McGraw Hill, 1989. 287 pages
Gottlieb, Irving M.,
Power Supplies, Switching Regulators, Inverters, and Converters, 2nd Edition, TAB Books, 1994. 479 pages.
Kassakian, John G., Martin F. Schlect, and George C. Verghese,
Principles of Power Electronics, Addison Wesley, 1991. 738 pages.
Lee, Yim Shu,
Computer Aided Analysis and Design of Switch Mode Power Supplies, Marcel Dekker, Inc., NY, 1993
Lenk, John D.,
Simplified Design of Switching Power Supplies, EDN Series for Design Engineers, Butterworth Heinmann, MA,
1994. 221 pages.
McLyman, C. W. T.,
Designing Magnetic Components for High Frequency DC DC Converters, KG Magnetics, San Marino, CA, 1993.
433 pages, 146 figures, 32 tables
Mitchell, Daniel,
Small Signal MathCAD Design Aids, e/j Bloom Associates, 115 Duran Drive, San Rafael, Ca 94903 2317,
415 492 8443, 1992. Computer disk included.
Mohan, Ned, Tore M. Undeland, William P. Robbins,
Power Electronics: Converter, Applications and Design, 2nd Edition, Wiley, 1995. 802 pages
Paice, Derek A.,
Power Electronic Converter Harmonics, Multipulse Methods for Clean Power, IEEE Press, 1995. 224 pages.
Whittington, H. W.,
Switched Mode Power Supplies: Design and Construction, 2nd Edition, Wiley, 1996 224 pages.
Basso, Christophe,
Switch Mode Power Supply SPICE Cookbook, McGraw Hill, 2001. CD ROM included. 255 pages.
http://onsemi.com
58
SMPSRM
Web Locations for Switching Mode Power Supply Information
Ardem Associates (Dr. R. David Middlebrook)
http://www.ardem.com/
Applied Power Electronics Conference (APEC)
The power electronics conference for the practical aspects of power supplies.
http://www.apec conf.org/
Dr. Vincent G. Bello s Home Page
SPICE simulation for switching mode power supplies.
http://www.SpiceSim.com/
e/j BLOOM Associates
(Ed Bloom) Educational Materials & Services for Power Electronics.
http://www.ejbloom.com/
The Darnell Group
(Jeff Shepard) Contains an excellent list of power electronics websites, an extensive list of manufacturer s contact
information and more.
http://www.darnell.com/
Switching Mode Power Supply Design by Jerrold Foutz
An excellent location for switching mode power supply information and links to other sources.
http://www.smpstech.com/
Institute of Electrical and Electronics Engineers (IEEE)
http://www.ieee.org/
IEEE Power Electronics Society
http://www.pels.org/pels.html
Power Control and Intelligent Motion (PCIM)
Articles from present and past issues.
http://www.pcim.com/
Power Corner
Frank Greenhalgh s Power Corner in EDTN
http://fgl.com/power1.htm
Power Designers
http://www.powerdesigners.com/
Power Quality Assurance Magazine
Articles from present and past issues.
http://powerquality.com/
Power Sources Manufacturers Association
A trade organization for the power sources industry.
http://www.psma.com/
Quantum Power Labs
An excellent hypertext linked glossary of power electronics terms.
http://www.quantumpower.com/
Ridley Engineering, Inc.
Dr. Ray Ridley
http://www.ridleyengineering.com/
http://onsemi.com
59
SMPSRM
Web Locations for Switching Mode Power Supply Information
(continued)
Springtime Enterprises Rudy Severns
Rudy Severns has over 40 years of experience in switching mode power supply design and static power conversion
for design engineers.
http://www.rudyseverns.com/
TESLAco
Dr. Slobodan Cuk is both chairman of TESLAco and head of the Caltech Power Electronics Group.
http://www.teslaco.com/
Venable Industries
http://www.venableind.com/
http://onsemi.com
60
SMPSRM
Analog ICs for SWITCHMODE Power Supplies
A number of different analog circuits that can be used for designing switchmode power supplies can be found in our
Analog IC Family Tree and Selector Guide (SGD504/D) available on our website at www.onsemi.com or through our
Literature Distribution Center, 800 344 3860 or (+1) 303 675 2175 or by email at ONlit@hibbertco.com. These
circuits are the same as those in the Power Management and System Management sections of the ON Semiconductor
Master Components Selector Guide, also available as SG388/D. Circuits used specifically for the off line controllers
and power factor controllers are in the Power Management section. Additional circuits that are frequently used with a
SMPS design (dc dc converters, voltage references, voltage regulators, MOSFET/IGBT drivers and dedicated power
management controllers) are included for reference purposes. Undervoltage and overvoltage supervisory circuits are in
the System Management section.
Information about the discrete semiconductors that are shown in this brochure and other discrete products that may
be required for a switching power supply can also be found in the ON Semiconductor Master Components Selector Guide
(SG388/D).
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61
SMPSRM
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SMPSRM
ON SEMICONDUCTOR STANDARD DOCUMENT TYPE DEFINITIONS
DATA SHEET CLASSIFICATIONS
A Data Sheet is the fundamental publication for each individual product/device, or series of products/devices, containing detailed
parametric information and any other key information needed in using, designing in or purchasing of the product(s)/device(s) it
describes. Below are the three classifications of Data Sheet: Product Preview; Advance Information; and Fully Released Technical Data.
PRODUCT PREVIEW
A Product Preview is a summary document for a product/device under consideration or in the early stages of development.
The Product Preview exists only until an Advance Information document is published that replaces it. The Product Preview
is often used as the first section or chapter in a corresponding reference manual. The Product Preview displays the following
disclaimer at the bottom of the first page: This document contains information on a product under development.
ON Semiconductor reserves the right to change or discontinue this product without notice.
ADVANCE INFORMATION
The Advance Information document is for a device that is NOT fully qualified, but is in the final stages of the release
process, and for which production is eminent. While the commitment has been made to produce the device, final
characterization and qualification may not be complete. The Advance Information document is replaced with the Fully
Released Technical Data document once the device/part becomes fully qualified. The Advance Information document
displays the following disclaimer at the bottom of the first page: This document contains information on a new product.
Specifications and information herein are subject to change without notice.
FULLY RELEASED TECHNICAL DATA
The Fully Released Technical Data document is for a product/device that is in full production (i.e., fully released). It
replaces the Advance Information document and represents a part that is fully qualified. The Fully Released Technical Data
document is virtually the same document as the Product Preview and the Advance Information document with the exception
that it provides information that is unavailable for a product in the early phases of development, such as complete parametric
characterization data. The Fully Released Technical Data document is also a more comprehensive document than either of
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DATA BOOK
A Data Book is a publication that contains primarily a collection of Data Sheets, general family and/or parametric information,
Application Notes and any other information needed as reference or support material for the Data Sheets. It may also contain cross
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APPLICATION NOTE
An Application Note is a document that contains real world application information about how a specific ON Semiconductor
device/product is used, or information that is pertinent to its use. It is designed to address a particular technical issue. Parts and/or
software must already exist and be available.
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A Selector Guide is a document published, generally at set intervals, that contains key line item, device specific information for
particular products or families. The Selector Guide is designed to be a quick reference tool that will assist a customer in determining
the availability of a particular device, along with its key parameters and available packaging options. In essence, it allows a customer
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The Master Components Selector Guide (SG388/D) is a listing of ALL currently available ON Semiconductor devices.
REFERENCE MANUAL
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